Chopper-stabilized instrumentation amplifier for impedance measurement

ABSTRACT

This disclosure describes a chopper stabilized instrumentation amplifier. The amplifier is configured to achieve stable measurements at low frequency with very low power consumption. The instrumentation amplifier uses a differential architecture and a mixer amplifier to substantially eliminate noise and offset from an output signal produced by the amplifier. Dynamic limitations, i.e., glitching, that result from chopper stabilization at low power are substantially eliminated through a combination of chopping at low impedance nodes within the mixer amplifier and feedback. The signal path of the amplifier operates as a continuous time system, providing minimal aliasing of noise or external signals entering the signal pathway at the chop frequency or its harmonics. The amplifier can be used in a low power system, such as an implantable medical device, to provide a stable, low-noise output signal.

This application is a continuation of U.S. patent application Ser. No.12/058,066, filed Mar. 28, 2008, which is a continuation of U.S. patentapplication Ser. No. 11/700,405, filed Jan. 31, 2007, now U.S. Pat. No.7,391,257, the entire content of each of which is incorporated herein byreference.

TECHNICAL FIELD

The invention relates to amplifiers and, more particularly, toinstrumentation amplifiers for accurate signal measurement.

BACKGROUND

Instrumentation amplifiers are used to accurately measure a variety oftest and measurement signals. A medical instrumentation amplifier, forexample, may be configured to measure physiological signals, such aselectrocardiogram (ECG), electromyogram (EMG), electroencephalogram(EEG), pressure, impedance, and motion signals. Typically,instrumentation amplifiers are constructed as differential amplifiersexhibiting low offset, low drift, low noise, high common mode rejection,high loop gain, and high input impedance. In many cases, instrumentationamplifiers may require careful matching and trimming of circuitcomponents to achieve a high degree of accuracy.

An instrumentation amplifier may be constructed with a discrete timeswitched capacitor architecture that obtains discrete signal samples.However, a discrete time architecture can produce undesirable aliasingof noise and signals, undermining the accuracy of measurement signals.Alternatively, an instrumentation amplifier may employ a chopperstabilized architecture in which a chopper circuit up-modulates ameasurement signal into a higher frequency band to remove noise andoffset. A chopper-stabilized architecture may have a limited bandwidth,however, producing a large ripple in the passband. The ripple may makeimplementation of chopper-stabilized designs difficult in low powerapplications.

SUMMARY

This disclosure describes a chopper stabilized instrumentationamplifier. The instrumentation amplifier is configured to achieve stablemeasurements at low frequency with very low power. The instrumentationamplifier uses a differential architecture and a mixer amplifier tosubstantially eliminate noise and offset from an output signal producedby the amplifier. Dynamic limitations, i.e., glitching, that result fromchopper stabilization at low power are substantially eliminated orreduced through a combination of chopping at low impedance nodes withinthe mixer amplifier and feedback. The signal path of the instrumentationamplifier operates as a continuous time system, providing minimalaliasing of noise or external signals entering the signal pathway at thechop frequency or its harmonics. In this manner, the instrumentationamplifier can be used in a low power system, such as an implantablemedical device, to provide a stable, low-noise output signal. Thechopper stabilized instrumentation amplifier may be used forphysiological signal sensing, impedance sensing, telemetry or other testand measurement applications.

In one embodiment, the invention provides a biomedical impedance sensingdevice comprising an alternating current (ac) source that generates anac stimulation current at a clock frequency for application to abiological load, a mixer amplifier coupled to receive a differentialinput signal from the load in response to the stimulation current,wherein the mixer amplifier amplifies the differential input signal toproduce an amplified signal and demodulates the amplified signal at theclock frequency to produce an output signal, a second modulator thatmodulates an amplitude of the output signal at the clock frequency, anda feedback path that applies the modulated output signal as adifferential feedback signal to the differential input signal.

In another embodiment, the invention provides a method comprisinggenerating an alternating current (ac) stimulation current at a clockfrequency, applying the stimulation current to a load to produce adifferential input signal, amplifying the differential input signal in amixer amplifier to produce an amplified signal, demodulating theamplified signal in the mixer amplifier at the clock frequency toproduce an output signal, modulating an amplitude of the output signalat the clock frequency to produce a differential feedback signal, andapplying the modulated output signal as a differential feedback signalto the differential input signal via a first feedback path.

In an additional embodiment, the invention provides a chopper-stabilizedinstrumentation amplifier comprising an alternating current (ac) sourcethat generates an ac stimulation current at a clock frequency forapplication to a load, a mixer amplifier coupled to receive adifferential input signal from the load in response to the stimulationcurrent, wherein the mixer amplifier amplifies the differential inputsignal to produce an amplified signal and demodulates the amplifiedsignal at the clock frequency to produce an output signal, a secondmodulator that modulates an amplitude of the output signal at the clockfrequency, and a feedback path that applies the modulated output signalas a differential feedback signal to the differential input signal.

The details of one or more embodiments of the invention are set forth inthe accompanying drawings and the description below. Other features,objects, and advantages of the invention will be apparent from thedescription and drawings, and from the claims.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram illustrating a chopper-stabilizedinstrumentation amplifier configured to achieve stable measurement atlow frequency with very low power.

FIG. 2 is a diagram illustrating a signal flow path of theinstrumentation amplifier of FIG. 1.

FIGS. 3A-D are graphs illustrating frequency components of a signal atvarious stages within the signal flow path of FIG. 2.

FIGS. 4A-D are graphs illustrating a signal at different stages withinthe signal flow path of FIG. 2.

FIG. 5 is graph illustrating exemplary noise performance of achopper-stabilized instrumentation amplifier.

FIG. 6 is a circuit diagram illustrating a chopper-stabilized mixeramplifier forming part of an instrumentation amplifier.

FIG. 7 is a block diagram illustrating an example embodiment of theinstrumentation amplifier of FIG. 1 in greater detail.

FIG. 8 is a circuit diagram illustrating an example embodiment of theinstrumentation amplifier of FIG. 1 for measurement of voltage signals.

FIG. 9 is a circuit diagram illustrating another example embodiment ofthe instrumentation amplifier of FIG. 1 for measurement of impedance.

FIG. 10 is a diagram illustrating a signal path flow for aninstrumentation amplifier in accordance with an embodiment of theinvention that includes a negative feedback path for constructing a highpass filter.

FIG. 11 is a circuit diagram illustrating the instrumentation amplifierof FIG. 10.

FIG. 12 is a diagram illustrating a signal path flow for aninstrumentation amplifier in accordance with an embodiment of theinvention that includes a positive feedback path for increasing inputimpedance.

FIG. 13 is a circuit diagram illustrating the instrumentation amplifierof FIG. 12.

FIG. 14A is a diagram illustrating a signal flow path for aninstrumentation amplifier in accordance with an embodiment of theinvention that is used to demodulate received telemetry signals.

FIG. 14B is a circuit diagram illustrating antenna input and feedbackcircuitry for the telemetry-configured instrumentation amplifier of FIG.14A.

FIG. 15A is a block diagram illustrating the telemetry-configuredinstrumentation amplifier of FIG. 14A.

FIG. 15B is a block diagram illustrating a clock synchronizer in FIG.15A in greater detail.

FIG. 16 is a block diagram illustrating an implantable medical deviceincluding one or more instrumentation amplifiers for measurement and/ortelemetry.

FIG. 17 is a block diagram illustrating a medical device programmerincluding one or more instrumentation amplifiers for telemetry.

DETAILED DESCRIPTION

This disclosure describes a chopper-stabilized instrumentationamplifier. The instrumentation amplifier is configured to achieve stablemeasurements at low frequency with very low power. The instrumentationamplifier uses a differential architecture and a mixer amplifier tosubstantially eliminate noise and offset from an output signal producedby the amplifier. Dynamic limitations, i.e., glitching, that result fromchopper stabilization at low power are substantially eliminated througha combination of chopping at low impedance nodes within the mixeramplifier and feedback. The signal path of the instrumentation amplifieroperates as a continuous time system, providing minimal aliasing ofnoise or external signals entering the signal pathway at the chopfrequency or its harmonics. In this manner, the instrumentationamplifier can be used in a low power system, such as an implantablemedical device, to provide a stable, low-noise output signal.

The chopper-stabilized instrumentation amplifier may be configured as amedical instrumentation amplifier, for example, to measure physiologicalsignals, such as electrocardiogram (ECG), electromyogram (EMG),electroencephalogram (EEG), pressure, impedance, motion signals, andother signals. In some embodiments, the instrumentation amplifier mayinclude a capacitor-based front end that is chopped to obtain lowfrequency voltage signals. In other embodiments, the instrumentationamplifier may include a current source-based front end that is choppedto obtain impedance measurements. In additional embodiments, theinstrumentation amplifier may include an antenna-based front end toobtain telemetry signals from other devices. The instrumentationamplifier may be useful not only in biomedical measurement applications,but also in general purpose test and measurement applications andwireless telemetry applications.

In general, an instrumentation amplifier, as described in thisdisclosure, may be configured for very low power applications. Animplantable medical device, for example, may be characterized by finitepower resources that are required to last several months or years.Accordingly, to promote device longevity, sensing and therapy circuitsare generally designed to consume very small levels of power. As anexample, operation of a sensor circuit incorporating an instrumentationamplifier, as described in this disclosure, may require a supply currentof less than 2.0 microamps, and more preferably less than 1.0 microamps.In some embodiments, such a sensor circuit may consume supply current ina range of approximately 100 nanoamps to 1.0 microamps. Such a sensormay generally be referred to as a micropower sensor. Although medicaldevices are described for purposes of illustration, a micropower sensormay be used in a variety of medical and non-medical test and measurementapplications. In each case, a sensor may be required to draw very lowpower, yet provide precise and accurate measurement.

According to various embodiments of this disclosure, achopper-stabilized instrumentation amplifier may include a front end, afirst chopper, an AC amplifier, a second chopper, an integrator in theform of a baseband amplifier with high gain and compensation, and atleast one feedback path. The amplifier, second chopper, and integratormay be referred to collectively as a mixer amplifier. The signal path ofthe instrumentation amplifier operates as a continuous time system,reducing aliasing of noise or other undesirable signals entering thesignal pathway at the chop frequency or its harmonics. The front endgenerates a differential input signal in the baseband, i.e., thefrequency band of interest for purposes of the test or measurementapplication. The baseband also may be referred to as the measurementband.

Amplification of the input signal can introduce DC offset and lowfrequency noise, such as 1/f or popcorn noise, due to amplifierimperfection or other factors. To reduce DC offset and low frequencynoise, a first chopper stage in the front end modulates the input signalat a chopper frequency prior to application of the input signal to themixer amplifier. After the input signal is amplified, the second chopperwithin the mixer amplifier demodulates the input signal at the chopperfrequency to produce an amplified output signal in the baseband. Thisprocess confines the noise and offset generated by the amplifier to thechopper frequency band, thereby preventing it from entering themeasurement band.

The mixer amplifier may have a modified folded cascode amplifierarchitecture in which the signal is chopped at low impedance nodes toprovide fast modulation dynamics. The mixer amplifier substantiallyremoves the noise and offset at the chopper frequency from thedemodulated signal, and thereby passes a low noise signal to themeasurement band. When the mixer amplifier is operating at low power,however, the bandwidth of the amplifier can be limited. Limitedbandwidth can result in glitching, i.e., ripple or spikes, in the outputsignal. An instrumentation amplifier as described in this disclosure mayprovide negative feedback to keep the signal change at the input to themixer amplifier relatively small. In addition, the feedback can beprovided to both inputs of the mixer amplifier to providedifferential-to-single conversion. As a result, an instrumentationamplifier can be configured to achieve a stable, low noise output whiledrawing very low current from a power source.

Additional feedback paths may be added to achieve increased performance.For example, a positive feedback path may used to increase inputimpedance of the instrumentation amplifier. As another example, anothernegative feedback path may allow for the construction of a high passfilter. Each feedback path may be a differential feedback path. Theseadditional feedback paths may not be necessary for the chopperstabilized amplifier to operate properly, but may enhance performance.For example, these feedback paths may be added to provide additionalsignal processing or conditioning that may be useful in variousapplications in which the instrumentation amplifier may be used.

Various example embodiments are presented. According to one exampleembodiment, which is useful when the instrumentation amplifier senses adifference in voltage across its inputs, the front end may include acontinuous time switched capacitor network. The switched capacitornetwork includes a differential set of switched input capacitors thattoggle between input voltages at a chop frequency. By chopping theswitched input capacitors, the input differential signal is up-modulatedto a chopper frequency, yielding a modulated signal at the differentialinput of the mixer amplifier. This embodiment may be useful as aninstrumentation amplifier for electroencephalography (EEG) andphysiological monitoring applications such as posture and activitymonitoring with accelerometers, catheter monitoring with pressuresensors, other pressure-related physiological monitoring, monitoring ofheart sounds, monitoring of brain signals, and other physiologicalmonitoring applications that require micro power systems for precisionsensor measurements.

According to another example embodiment, the instrumentation amplifiermay be configured to measure impedances of physiologic importance, suchas tissue impedance Measuring such impedances can be used to measurephysiological conditions, such as pulmonary edema, minute ventilationrespiration (e.g., for sleep apnea), cardiac dynamics, and generaltissue impedance. It is important when measuring such impedances thatthe stimulation current be small, e.g., less than or equal toapproximately 10 μA or less, to avoid stimulation of excitable cells, orcause other detrimental effects such as electrode corrosion. In thisexample embodiment, the front end produces an AC modulated signal thatis AC coupled to the mixer amplifier through tissue of a patient. Thefront end modulates a stimulation current at the chopper frequency tomodulate the amplitude of a tissue voltage signal in response to thestimulation current. In this way, the tissue is not exposed to DCcurrent. The relative phase between the clock driving the stimulationcurrent and the clock driving the chop frequency of the mixer amplifiercan be changed to allow the instrumentation amplifier to measure eitherthe resistance or reactance of the tissue. For resistance, the chopfrequencies of the front end and the mixer amplifier ordinarily will bein-phase with one another.

According to an additional example embodiment, the instrumentationamplifier may be configured to be useful in telemetry applications,e.g., as a down mixer in a receiver. In this example embodiment, theinstrumentation amplifier may be located in a patient or clinicianprogrammer or an implantable pulse generator (IPG) or other implantablemedical device (IMD) implanted within a patient that communicates, viawireless radio frequency (RF) telemetry, with the clinician or patientprogrammer. The front end in this example embodiment includes atransmitter located in a remote transmitting device, and a receiveantenna in the receiving device for receiving a telemetry signal fromthe transmitter. The telemetry signal may, for example, have a frequencyin a range of approximately 10 kHz to 1 GHz, and in some embodimentsapproximately 175 kHz, although other frequencies are possible. In thisexample, the first chopper actually resides in the transmitter of theremote device. The front end couples the transmitted signal, which is asignal modulated at the chopper frequency, to the mixer amplifier whichdirectly down-modulates the signal to baseband while substantiallyeliminating 1/f noise and offset from the mixer. A phase locked loop, orother clock synchronization circuit, may be included to provide feedbackto keep the transmitter (front end) and receiver (mixer amplifier) inphase with each other.

Telemetry signals may include data, programming instructions of thelike. For example, a medical device programmer may transmit telemetrysignals to an implanted medical device to download programminginstructions that alter operational aspects of the implanted medicaldevice, such as therapies delivered by the implanted medical device. Theprogramming instructions may specify new stimulation or drug deliveryprograms or adjustments to existing programs. The programminginstructions may specify adjustments to programming parameters, such aselectrical stimulation pulse amplitude, pulse width, pulse rate, orduration, or drug delivery dosage, drug delivery rate, dosage limits,lockout intervals, or the like. Likewise, an implanted medical devicemay transmit data to an external programmer via the telemetry signals.The data may transmitted to the programmer may include operational data,diagnostic data, fault data, sensor data, or the like.

Physiological signals are generally found at low frequencies, e.g., lessthan or equal to approximately 100 Hz and, in many cases, less than orequal to approximately 2 Hz, or less than or equal to approximately 1Hz. Measurement and analysis of physiological signals can be used todiagnose chronic or acute disease states and other medical conditions.Example physiological signals include EEG signals, ECG signals, EMGsignals, pressure, impedance, and motion signals, as previouslydescribed. Such signals may be used to detect or measure cardiacischemia, pulmonary edema, breathing, activity, posture, pressure, brainactivity, gastrointestinal activity, and the like.

Implantable medical devices including instrumentation amplifiers used tomeasure such physiological signals may be required to operate with lownoise and low power. Low power consumption may be especially importantin chronically implanted medical devices designed for several years ofservices, and particularly those medical devices configured to sensephysiological signals and deliver therapies. Examples of therapeuticmedical devices are implantable cardiac pacemakers, implantablecardioverter-defibrillators, implantable electrical stimulators, such asneurostimulators, muscle stimulators or other tissue stimulators,implantable drug delivery devices, and other devices.

It is important that an instrumentation amplifier provide low noiseperformance so that noise does not result in reduced sensitivity orwrong or misleading diagnostic information. It is also important thatthe instrumentation amplifier operate with low power in order toconserve limited battery resources and thereby promote operationallongevity of the implantable medical device. A chopper-stabilizedinstrumentation amplifier, as described in this disclosure, may beconfigured to achieve precise measurements at low frequency with lowpower. As will be described, a chopper-stabilized instrumentationamplifier can be configured to apply chopping at low impedance nodes andapply feedback to reduce ripple resulting from low bandwidth of theamplifier.

FIG. 1 is a block diagram illustrating a chopper stabilizedinstrumentation amplifier 10 that is configured to achieve stablemeasurement at low frequency with very low power. Instrumentationamplifier 10 uses a differential architecture and a mixer amplifier tosubstantially eliminate 1/f noise, popcorn noise, and offset. Dynamiclimitations, i.e., glitching, that result from chopper stabilization atlow power are eliminated through a combination of chopping at lowimpedance nodes within a mixer amplifier 14 and feedback via feedbackpath 16. The signal path of the instrumentation amplifier operates as acontinuous time system, providing minimal aliasing of noise or externalsignals entering the signal pathway at the chop frequency or itsharmonics. As a result, instrumentation amplifier 10 can provide stablemeasurements for low frequency signals, such as physiological signalsand other signals having a frequency of less than approximately 100 Hz,and preferably less than or equal to approximately 2.0 Hz, and morepreferably less than or equal to approximately 1.0 Hz, while operatingunder the constraints of a micro power system, e.g., drawing a supplycurrent of less than or equal to approximately 2.0 microamps, and morepreferably less than or equal to approximately 1.0 microamps, andrequiring a supply voltage of less than or equal to approximately 2.0volts, and more preferably less than or equal to approximately 1.5volts.

As shown in FIG. 1, instrumentation amplifier 10 includes front end 12,mixer amplifier 14, and feedback path 16. In the example of FIG. 1,front end 12 may provide a switched or static capacitive differentialinterface to mixer amplifier 14, e.g., for measurement of a lowfrequency voltage amplitude. In other embodiments, front end 12 may beconfigured for impedance measurement or telemetry applications. Frontend 12 couples a differential modulated (chopped) input signal thatcarries a low frequency signal of interest on a carrier (chopper)frequency. In other words, front end 12 shifts a low frequency signalthat is subject to introduction of low frequency noise by mixeramplifier 14 to a carrier frequency at which the mixer amplifier 14 doesnot introduce substantial noise into the signal. The low frequencysignal of interest may have, for example, a frequency within a range of0 to approximately 100 Hz. In some embodiments, the carrier (chopper)frequency may be within a frequency range of approximately 4 kHz to 200kHz. Front end 12 modulates the low frequency signal prior tointroduction to mixer amplifier 14 so that the original baseband (lowfrequency) signal components are not corrupted by noise componentsintroduced by mixer amplifier 14 at low frequency.

Noise generally enters the signal path of instrumentation amplifier 10through mixer amplifier 14. However, mixer amplifier 14 should notintroduce noise to the modulated signal at the carrier frequency.Rather, the noise components are typically present at low frequency andmay include 1/f noise or popcorn noise. In addition, noise in the formof dc offset cannot be introduced at the carrier frequency. Mixeramplifier 14 receives and amplifies the up-modulated input signal fromfront end 12. Again, the up-modulated input signal is up-modulated tothe chopper frequency to protect the input signal from low frequencynoise and offset.

Mixer amplifier 14 demodulates the modulated input signal from thecarrier frequency to the baseband of interest while upmodulating themixer amp 1/f noise and offset out of the measurement band. Thus, theoriginal low frequency signal components are demodulated back tobaseband without the low frequency noise and offset components of themixer amplifier 14. Mixer amplifier 14 passes only the baseband signals,i.e., signals with frequency components of approximately 100 Hz or less,as output and substantially reduces or eliminates the noise componentslocated at the carrier frequency. Thus, the output of instrumentationamplifier 10 contains the low frequency signal components of interest.In addition, mixer amplifier 14 provides a gain amplifier that amplifiesthe input signal. In this way, instrumentation amplifier 10 provides alow noise output while operating at low power.

Instrumentation amplifier 10 operates under the constraints of a micropower system and therefore has limited bandwidth. The limited bandwidthof instrumentation amplifier 10 can cause glitching or ripple in thepassband of the output signal. As will be described, mixer amplifier 14may have a modified folded cascode architecture that provides switching,e.g., via CMOS switches, at low impedance nodes. Switching at lowimpedance nodes enables chopping at higher frequencies where the onlylimitation would be the charge injection residual offset.

Feedback path 16 is coupled between the output of mixer amp 14 and frontend 12 to reduce the ripple. Feedback path 16 may have a differentialconfiguration that substantially eliminates glitching in the outputsignal by driving the net input signal to mixer amplifier 14 towardzero. In this way, feedback path 16 keeps the signal change at the inputof mixer amplifier 14 relatively small in steady state. As a result,instrumentation amplifier 10 achieves a stable, low noise, lowdistortion output while operating at low power.

Instrumentation amplifier 10 may be useful in many differentapplications. This disclosure presents various example embodiments ofinstrumentation amplifier 10. However, these example embodiments shouldnot be considered limiting of the instrumentation amplifier 10 asbroadly embodied and described in this disclosure. Rather, it should beunderstood that the example embodiments described in this disclosure area subset of many different example embodiments within the scope of thisdisclosure.

In some embodiments, a device such as an implantable medical device mayinclude multiple instrumentation amplifiers 10. For example, multipleinstrumentation amplifiers 10 may be fabricated in parallel to providemultiple sensing channels. The multiple sensing channels may sense thesame type of physiological information, e.g., at different positions orangles, or via different sensors. In addition, multiple sensing channelsmay sense different types of physiological information, such asimpedance, ECG, EEG, EMG, pressure, motion, and the like.

According to one example embodiment, front end 12 of amplifier 10 maycomprise a continuous time switched capacitor network. The switchedcapacitor network includes a differential set of switched inputcapacitors that toggle between input voltages at the positive andnegative terminals of instrumentation amplifier 10. By toggling theswitched input capacitors at the chopper frequency, the differentialinput signal is chopped. In this manner, the differential input signalis up-modulated to the carrier frequency, yielding a modulated signal atthe differential input of mixer amplifier 14. In this example,instrumentation amplifier 10 may be implemented to measure physiologicalvoltage signals such as ECG, EEG, EMG, pressure, motion, or the like.Accordingly, the inputs to front end 12 may be electrodes, or outputsfrom any of a variety of accelerometers, pressure sensors, strain gaugesensors, or the like.

According to another example embodiment, front end 12 of instrumentationamplifier 10 may comprise an impedance sensor. In particular,instrumentation amplifier 10 may form a biological impedance sensingdevice for measuring the impedance of tissue of a patient, e.g., muscletissue, organ tissue, brain tissue, adipose tissue, or a combination oftissues. The impedance sensor formed by front end 12 produces an ACmodulated signal that is AC coupled to mixer amplifier 14 through thetissue of the patient. In this case, front end 12 modulates astimulation current to modulate the amplitude of a tissue voltagesignal. In other words, front end 12 chops the stimulation currentsource. Thus, the patient is not exposed to a direct current (DC)signal. Moreover, the modulated signal may not substantially excite thetissue, thereby decreasing the likelihood that the patient mayexperience discomfort or other detrimental effects from the modulatedsignal. The relative phase between the clock driving the stimulationcurrent and the clock driving the chop frequency of mixer amplifier 14can be changed to allow the instrumentation amplifier to measure eitherthe resistance or reactance of the tissue. Consequently, instrumentationamplifier 10 may be used to measure a variety of physiological signals,e.g., for pulmonary edema, minute ventilation (sleep apnea), cardiacdynamics, and general tissue impedance. For example, the relative phasebetween the stimulation current and mixer amplifier clocks may bedynamically adjusted to obtain different types of measurement, e.g.,resistance or reactance, during the course of measurement.

According to an additional example embodiment, feedback 16 may include asecond feedback path in addition to the previously described negativefeedback path that reduces glitching in the output of instrumentationamplifier 10 and provides the nominal gain for amplifier 10. This secondfeedback path provides negative feedback to allow for the constructionof a high pass filter. The second feedback path is dominant at lowfrequencies, i.e., frequencies lower than the cutoff frequency, and thechopper stabilized negative feedback path is dominant at passbandfrequencies. The high pass filter may have a cutoff frequencyapproximately equal to, e.g., approximately 2.5 Hz, or 0.5 Hz, or 0.05Hz. In this case, the first feedback path, i.e., the “chopperstabilizing” feedback path that eliminates glitching at the output, isdominant at pass band frequencies and the second “high-pass filter”feedback path is dominant at low frequencies. The corner frequency ofthe high pass filter in the second feedback path can be set by thescaling of feedback capacitors in the first feedback path and thetime-constant of a switched capacitor integrator in the second feedbackpath. As one example, the high pass filter provided by this feedbackpath may be useful for rejecting post-pacing artifacts in heartmonitoring applications and filtering out electrode offsets. The secondfeedback path may include a high-pass integrator that is chopperstabilized for the lowest 1/f noise floor.

According to yet another embodiment, feedback 16 may include a thirdfeedback path in addition to the first feedback path. The third feedbackpath provides positive feedback to increase the input impedance ofinstrumentation amplifier 10. The increased input impedance is achievedby sampling the output of instrumentation amplifier 10 and applying ascaled charge to the input of the switched capacitors in front end 12 toprovide compensatory charge at the sensor input. The scaled charge maybe applied at a point in the signal flow prior to chopping of the inputsignal. The injected current effectively “replaces” charge lost duringthe sampling of the input chopper capacitors in front end 12. Thischarge replacement feedback may be considered similar to base currentcompensation. The positive feedback may increase the equivalentlow-frequency input impedance of instrumentation amplifier 10 by anorder of magnitude or more. This third feedback path may not benecessary in various applications. If increased input impedance isdesired, however, this third feedback path can be readily added.

According to a further example embodiment, instrumentation amplifier 10may include the previously described second and third feedback paths inaddition to the first (chopper stabilizing) feedback path. In this case,the third feedback path does not tap off of the output signal ofinstrumentation amplifier 10 as previously described. Rather, the third,positive feedback path may tap off of an integrated signal provided bythe second, high-pass filter feedback path. Accordingly, variouscombinations of first, second, and/or third feedback paths may beprovided to address glitching, low frequency rejection, and/or amplifierinput impedance.

In another example embodiment, instrumentation amplifier 10 may be usedin telemetry applications and, more particularly, telemetry applicationsoperating at relatively low frequencies and low power, e.g., on theorder of approximately 175 kHz in a medical device. For example,instrumentation amplifier 10 may be used as a telemetry receiver in animplantable pulse generator (IPG), implantable drug pump, or otherimplantable medical device (IMD) implanted within a patient thatcommunicates, via wireless radio frequency (RF) telemetry, with aclinician or patient programmer, or with other implanted or externalmedical devices. Instrumentation amplifier 10 may also be used, in areciprocal manner, as a telemetry receiver in a clinician or patientprogrammer that communicates with an IPG implanted within a patient.When implemented as a telemetry receiver, front end 12 may include atransmitter and a receive antenna for receiving a transmitted signalfrom the transmitter. However, the transmitter portion of front end 12actually resides in the remote device that transmits the signal. Frontend 12 couples the received signal to mixer amplifier 14, whichdirectly-down mixes the received signal to baseband while substantiallyeliminating 1/f noise and offset. A phase locked loop may providefeedback to keep the clocks at the transmitter and receiver in phasewith each other.

Instrumentation amplifier 10 can provide one or more advantages in avariety of embodiments. For example, as previously described,instrumentation amplifier 10 can achieve stable measurements at lowfrequency with low power. This is a result of the basic architecture ofinstrumentation amplifier 10. As another advantage, on-chip, poly-polycapacitors may be used to implement feedback capacitors ininstrumentation amplifier 10. Poly-poly capacitors enable fast switchingdynamics and can be formed on-chip with other amplifier components. Apoly-poly capacitor may be formed on chip with other devices bycombining two polysilicon electrodes and an intervening silicon dioxidedielectric. The gain of the instrumentation amplifier can be set by theratio of the feedback capacitors to the input capacitors and centeredaround a selected reference voltage. Further, by modulating the inputsignal at front end 12, the common mode input voltage can swing fromrail to rail and mixer amplifier 14 is still able to extract adifferential voltage. These advantages are merely exemplary and shouldbe considered a subset of potential advantages provided byinstrumentation amplifier 10. Additional advantages are discussed inthis disclosure or may occur to those skilled in the art uponconsideration of this disclosure. Moreover, such advantages may notcoexist in every embodiment.

FIG. 2 is a block diagram illustrating a signal path flow of anexemplary instrumentation amplifier 10. In FIG. 2, front end 12 includesmodulator 20 for modulating a low frequency input signal 32 to producemodulated input signal 21. An input capacitance (Cin) 13 couples theoutput of modulator 20 to summing node 22. For a differential inputsignal, Cin 13 may include a first input capacitor coupled to a firstinput of mixer amplifier 14 and a second input capacitor coupled to asecond input of mixer amplifier 14. Modulator 20 modulates adifferential amplitude of input signal 32 to a carrier frequencyprovided by clock signal 21A. Clock signal 21A, like other clock signalsdescribed in this disclosure, may be a square wave signal thateffectively multiples the signal by plus 1 and minus 1 at a desiredclock frequency. In this manner, module 20 chops the input signal 32prior to application of the input signal to mixer amp 14. Modulator 20may, in some embodiments, comprise a pair of complementary metal oxidesemiconductor (CMOS) single pole, double throw (SPDT) switches that aredriven by clock signal 21A to modulate (chop) input signal 32 to thecarrier frequency. The CMOS SPDT switches may be cross-coupled to eachother to reject common mode signals.

In one example embodiment, the CMOS switches may be coupled to a set ofdifferential capacitors to form a continuous time switched capacitornetwork that forms input capacitance Cin at the input of mixer amplifier14. In this case, front end 12 may be coupled to a physiological sensorthat generates an input signal 32 proportional to a sensed physiologicalparameter at its outputs. For example, input signal 32 may be adifferential output signal from a pair or electrodes, or from anaccelerometer, pressure sensor, or the like. In another exampleembodiment, the CMOS switches may be coupled to capacitors that ACcouple modulated input signal 21 to the input of mixer amplifier 14. Inthis case, front end 12 may be an impedance sensor that modulates astimulation current which is applied across tissue of a patient. In anadditional embodiment, front end 12 may be part of a telemetrytransmitter. In this case, input signal 32 is an electrical signalencoded with data that is modulated to the carrier frequency by clocksignal 21A for transmission over a wireless channel.

Feedback summing node 22 will be described below in conjunction withfeedback path 16. Summing node 24 represents the introduction of offsetand 1/f noise within mixer amplifier 14. At summing node 24, theoriginal baseband components of input signal 32 are located at thecarrier frequency. The baseband signal components of input signal 32 mayhave a frequency within a range of 0 to approximately 100 Hz and thecarrier frequency may be approximately 4 kHz to approximately 10 kHz.Noise 23 enters the signal pathway at summing node 24 to produce noisymodulated input signal 25. Noise 23 may include 1/f noise, popcornnoise, offset, and any other external signals that may enter the signalpathway at low (baseband) frequency. At node 24, however, the originallow frequency components have already been chopped to a higher frequencyband by modulator 20. Thus, the low frequency noise 23 is segregatedfrom the original low frequency components.

Mixer amplifier 14 receives noisy modulated input signal 25 from node24. In the example of FIG. 2, mixer amplifier 14 includes gain amplifier26, modulator 28, and integrator 30. Amplifier 26 amplifies noisymodulated input signal 25 to produce amplified signal 27. Modulator 28demodulates amplified signal 27. That is, modulator 28 modulates noise23 up to the carrier frequency and demodulates the original basebandsignal components from the carrier frequency back to baseband. Modulator28 may comprise switches, e.g., CMOS SPDT switches, located at lowimpedance nodes within a folded-cascode architecture of mixer amplifier14. Modulator 28 is supplied with clock signal 21B to demodulateamplified signal 27 at the same carrier frequency as clock signal 21A.Hence, clock signals 21A, 21B should be synchronous with each other. Insome embodiments, clock signal 21A and clock signal 21B may be the samesignal, i.e., supplied by the same clock. In other embodiments, e.g.,for measurement of reactance, the relative phasing of clock signals 21A,21B and 21C may be altered.

In some embodiments, clock signal 21A and clock signal 21B may besupplied by different clocks. In such embodiments, modulators 20 and 28may not be precisely in phase with each other and additional circuitrymay be added to ensure that clock signals 21A and 21B remain in phasewith each other. This is the case when instrumentation amplifier 10 isused as a telemetry receiver because modulator 20 may be used by atransmitter in a remote device to modulate the signal for transmissionover a wireless channel while modulator 28 may be used by the receiverto demodulate the received signal. Thus, additional signal processing,such as a phase locked loop, may be used to keep modulators 20, 28 inphase with each other.

Integrator 30 operates on demodulated signal 29 to pass the lowfrequency signal components at baseband and substantially eliminatenoise components 23 at the carrier frequency. In this manner, integrator30 provides compensation and filtering. In other embodiments,compensation and filtering may be provided by other circuitry. However,the use of integrator 30 as described in this disclosure may bedesirable. FIG. 6 provides a detailed circuit diagram of an exampleembodiment of mixer amplifier 14. Feedback path 16, as shown in FIG. 2,provides negative feedback to the input of mixer amp 14 to reduceglitching in output signal 31. In particular, feedback path 16 drivesmodulated signal 25 toward zero in steady state. In this way, feedback16 keeps the signal change at the input to mixer amplifier 14 small.Feedback path 16 includes a modulator 34, which modulates output signal31 to produce a differential feedback signal 35 that is added to thesignal path between front end 12 and mixer amplifier 14 at node 22.

Feedback path 16 provides capacitor scaling versus the input capacitanceCin of mixer amplifier 14 to produce attenuation and thereby generategain at the output of amplifier 10. Accordingly, feedback path 16 mayinclude a feedback capacitance (Cfb) 17 that is selected to producedesired gain, given the value of the input capacitance (Cin) 13 of mixeramplifier 14. Integrator 30 may be designed to provide a stable feedbackpath 16 with acceptable bandwidth while also filtering out theupmodulated offset and 1/f noise from the measurement band.

Clock signal 21C drives modulator 34 in feedback path 16 to modulateoutput signal 31 at the carrier frequency. Clock signal 21C may bederived from the same clock as clock signal 21B. However, because outputsignal 31 is single ended, feedback 16 includes two feedback paths thatapply the negative feedback to the positive and negative input terminalsof mixer amplifier 14. Thus, the two feedback paths should be 180degrees out of phase with each other, with one of the feedback pathsmodulating synchronously with modulator 28. This ensures that a negativefeedback path exists during each half of the clock cycle.

As an alternative, in some embodiments, mixer amplifier 14 may beconfigured to generate a differential output signal, rather than asingle-ended output signal. A differential output signal may providepositive and negative outputs. In this case, feedback path 16 can feedback the positive output to the positive input of mixer amplifier 14 andfeed back the negative output to the negative input of the mixeramplifier. For a differential output signal, feedback path 16 wouldmodulate each of the positive and negative outputs. However, thepositive and negative outputs could be modulated in-phase, rather thanout of phase. Although a differential output is possible, a feedbackpath 16 configured to convert a single-ended output to differentialfeedback will be described herein for purposes of illustration.

In FIG. 2, only the previously described negative feedback path 16 isshown. That is, the previously described feedback paths for increasinginput impedance and constructing a high pass filter are excluded fromFIG. 2. These feedback paths are excluded in FIG. 2 because they are notnecessary for proper operation of instrumentation amplifier 10. Thefeedback paths, however, are included in the signal flow path diagramsin FIGS. 10 and 12, and may be highly desirable in some applications.

FIGS. 3A-3D are graphs illustrating the frequency components of a signalat various stages within the signal flow path of FIG. 2. In particular,FIG. 3A illustrates the frequency components of input signal 32. Thefrequency components are represented by block 40 and located at basebandin FIG. 3A.

FIG. 3B illustrates the frequency components of noisy modulated inputsignal 25. In FIG. 3B, the original baseband frequency components ofnoisy modulated input signal 25 are modulated and represented by blocks42 at the odd harmonics. The frequency components of noise 23 arerepresented by dotted line 43. It is clear in FIG. 3A that the energy ofthe frequency components of noise 23 is located at baseband and energyof the original low frequency components is located at the carrier(chop) frequency and its odd harmonics.

FIG. 3C illustrates the frequency components of demodulated signal 29.In particular, the original low frequency components of demodulatedsignal 29 are located back at baseband and represented by block 44. Thefrequency components of noise 23 are modulated and represented by dottedline 45. The frequency components of noise 23 are located at the carrier(chop) frequency odd harmonics in FIG. 3C. FIG. 3C also illustrates theeffect of a low pass filter that may be applied to demodulated signal 29by integrator 30. The low pass filter effect is represented by dashedline 49.

FIG. 3D is a graph that illustrates the frequency components of outputsignal 31. In FIG. 3D, the frequency components of the original lowfrequency components are represented by block 46 and the frequencycomponents of noise 23 are represented by dotted line 47. FIG. 3Dillustrates that integrator 30 removes the frequency components fromnoise 23 that were located outside of the passband of the low passfilter shown in FIG. 3C. Clearly, the energy from noise 23 issubstantially eliminated from output signal 31, or at leastsubstantially reduced relative to the original noise and offset thatotherwise would be introduced.

FIGS. 4A-4D are graphs illustrating the step response time domainbehavior of a chopper stabilized signal at different stages withininstrumentation amplifier 10. In particular, with reference to FIG. 2,FIGS. 4A-4D illustrate the time domain behavior of noisy modulated inputsignal 25, amplified signal 27, demodulated signal 29, and output signal31, respectively. For reference, each of FIGS. 4A-4D also illustratesignals 52, 54, 56, 58 and a selected reference voltage 50. Signals 52,54, 56, and 58 correspond to signals 25, 27, 29, and 31, respectively,and illustrate the time domain behavior without negative feedback viafeedback path 16. In FIGS. 4A-4C, signals 25, 27, and 29 are centeredaround reference voltage 50 at time zero, and suppressed towardreference voltage 50 over time by negative feedback. Hence, by addingnegative feedback via feedback path 16, ac signals are driven to zero insteady state.

In general, FIGS. 4A-4D illustrate elimination of transient glitcheswithin instrumentation amplifier 10 through the use of feedback path 16and switching at low impedance nodes within mixer amplifier 14. Thisglitching results from the dynamic limitations of instrumentationamplifier 10. However, feedback 16 substantially suppresses theglitching by driving the active signal within mixer amplifier 14 towardzero, or reference voltage 50 in FIGS. 4A-4D, in steady state.

The graph in FIG. 4A shows noisy modulated input signal 25 andcorresponding signal 52 without negative feedback. Signals 25 and 52 arecentered around reference voltage 50. Noisy modulated input signal 25 isamplified by mixer amplifier 14 to generate amplified signal 27.

As shown in FIG. 4B, the limited bandwidth of amplifier 26 tends tosoften or round the edges of amplified signal 27 and correspondingsignal 54 due to its finite rise time. When amplified signal 27 isdemodulated with a square wave, demodulated signal 29 appears as aseries of spikes superimposed on the desired signal, as shown in FIG.4C. Accordingly, output signal 31 also appears as a series of spikessuperimposed on the desired signal in FIG. 4D. The spikes in outputsignal 31 can create a significant sensitivity error because the spikessubtract energy from the desired signal. In addition, the spikes aredifficult to suppress to an acceptable level without a very high orderlow pass filter. Moreover, the spikes are particularly problematicbecause the spikes may be similar to signals that may be of interest,such as intrinsic and evoked ECG heart potentials or EEG seizureactivity.

Instrumentation amplifier 10 substantially suppresses the glitching insteady state through feedback 16. Feedback 16 applies output signal 31back to the input of mixer amplifier 14 to drive noisy modulated signal25 toward zero in steady state. Consequently, little dynamic performanceis required of mixer amplifier 14. This is achieved through partitioningthe modulation processes before the signal is integrated in mixeramplifier 14, which decouples the overall loop dynamics from theswitching (modulating) dynamics. Moreover, by closing the feedback path,the overall gain of instrumentation amplifier 10 is set by the ratio ofthe input capacitors, i.e., capacitors Cin in front end 12, and feedbackcapacitors, i.e., capacitors Cfb in feedback path 16. Setting gainthrough capacitors ratios makes sensitivity generally immune to processvariations in the transistors. In this way, feedback 16 enablesinstrumentation amplifier 10 to achieve stable (low-noise) measurementsat low frequency with very low power.

The gain of instrumentation amplifier 10 may be different for differentapplications. For ECG sensing, for example, a gain of approximately 50may be desirable. For EEG sensing, a gain closer to 500 may bedesirable. As one example, Cin could be set to 20 picofarads (pF) andCfb could be set to 40 femtofarads (fF) to achieve a gain ofapproximately 500, e.g., for EEG sensing. As another example, Cin couldbe set to 10 pF and Cfb could be set to 200 fF to achieve a gain ofapproximately 50.

FIG. 5 is a bode plot illustrating exemplary noise performance ofinstrumentation amplifier 10. In particular, lines 58 and 59 in the bodeplot represent the noise prior to chopping (prior to the input of mixeramplifier 14), and the noise after chopping (at the output of mixeramplifier 14), respectively. Line 58 shows that the noise content priorto chopping is primarily located at low frequency. At high frequency,only white noise is present. In a preferred embodiment, the chopfrequency is above the corner of the 1/f noise and thermal noiseintercept point. Accordingly, line 59 shows that the noise contained inthe signal after chopping is substantially eliminated. The noise that iscontained in the signal after chopping is essentially the theoreticalwhite noise limit.

FIG. 6 is a circuit diagram illustrating an example embodiment of mixeramplifier 14 of instrumentation amplifier 10 in greater detail. Aspreviously described, mixer amplifier 14 amplifies noisy modulated inputsignal 25 to produce an amplified signal and demodulates the amplifiedsignal. Mixer amplifier 14 also substantially eliminates noise from thedemodulated signal to generate output signal 31. In the example of FIG.6, mixer amplifier 14 is a modified folded-cascode amplifier withswitching at low impedance nodes. The modified folded-cascodearchitecture allows the currents to be partitioned to maximize noiseefficiency. In general, the folded cascode architecture is modified inFIG. 6 by adding two sets of switches. One set of switches isillustrated in FIG. 6 as switches 60A and 60B (collectively referred toas “switches 60”) and the other set of switches includes switches 62Aand 62B (collectively referred to as “switches 62”).

Switches 60 are driven by chop logic to support the chopping of theamplified signal for demodulation at the chop frequency. In particular,switches 60 demodulate the amplified signal and modulate front-endoffsets and 1/f noise. Switches 62 are embedded within a self-biasedcascode mirror formed by transistors M6, M7, M8 and M9, and are drivenby chop logic to up-modulate the low frequency errors from transistorsM8 and M9. Low frequency errors in transistors M6 and M7 are attenuatedby source degeneration from transistors M8 and M9. The output 31 ofamplifier 26 is at baseband, allowing an integrator formed by transistorM10 and capacitor 63 (Ccomp) to stabilize feedback path 16 (not shown inFIG. 6) and filter modulated offsets.

Mixer amplifier 14 has three main blocks: a transconductor, ademodulator, and an integrator. The core is similar to a folded cascode.In the transconductor section, transistor M5 is a current source for thedifferential pair of input transistors M1 and M2. In some embodiments,transistor M5 may pass approximately 800 nA, which is split betweentransistors M1 and M2, e.g., 400 nA each. Transistors M1 and M2 are theinputs to amplifier 14. Small voltage differences steer differentialcurrent into the drains of transistors M1 and M2 in a typicaldifferential pair way. Transistors M3 and M4 serve as low side currentsinks, and may each sink roughly 500 nA, which is a fixed, generallynonvarying current. Transistors M1, M2, M3, M4 and M5 together form adifferential transconductor.

In this example, approximately 100 nA of current is pulled through eachleg of the demodulator section. The AC current at the chop frequencyfrom transistors M1 and M2 also flows through the legs of thedemodulator. Switches 60 alternate the current back and forth betweenthe legs of the demodulator to demodulate the measurement signal back tobaseband, while the offsets from the transconductor are up-modulated tothe chopper frequency. As discussed previously, transistors M6, M7, M8and M9 form a self-biased cascode mirror, and make the signalsingle-ended before passing into the output integrator formed bytransistor M10 and capacitor 63 (Ccomp). Switches 62 placed within thecascode (M6-M9) upmodulate the low frequency errors from transistors M8and M9, while the low frequency errors of transistor M6 and transistorM7 are suppressed by the source degeneration they see from transistorsM8 and M9. Source degeneration also keeps errors from Bias N2transistors 66 suppressed. Bias N2 transistors M12 and M13 form a commongate amplifier that presents a low impedance to the chopper switchingand passes the signal current to transistors M6 and M7 with immunity tothe voltage on the drains.

The output DC signal current and the upmodulated error current pass tothe integrator, which is formed by transistor M10, capacitor 63, and thebottom NFET current source transistor M11. Again, this integrator servesto both stabilize the feedback path and filter out the upmodulated errorsources. The bias for transistor M10 may be approximately 100 nA, and isscaled compared to transistor M8. The bias for lowside NFET M11 may alsobe approximately 100 nA (sink). As a result, the integrator is balancedwith no signal. If more current drive is desired, current in theintegration tail can be increased appropriately using standard integratecircuit design techniques. Various transistors in the example of FIG. 6may be field effect transistors (FETs), and more particularly CMOStransistors.

FIG. 7 is a block diagram illustrating instrumentation amplifier 10 ingreater detail. It should be understood that FIG. 7 is merely exemplaryand should not be considered limiting of the invention as described inthis disclosure in any way. Rather, it is the purpose of FIG. 7 toprovide an overview that is used to describe the operation ofinstrumentation amplifier 10 in greater detail. This overview is used asa framework for describing the previously mentioned example embodimentswith respect to the detailed circuit diagrams provided in thisdisclosure.

In FIG. 7, front end 12 outputs a modulated differential input signal25. The modulated differential input signal carries the signal ofinterest at a carrier frequency. As previously described, front end 12may take the form of various different components. Front end 12 may, forexample, be a continuous time switched capacitor network that modulates(chops) an input signal from a physiological sensor, an impedance sensorthat modulates a stimulation current to produce an AC modulated signalthat is AC coupled to mixer amplifier 14 through tissue of a patient, orpart of a telemetry transmitter that modulates the data encoded outputsignal to a carrier frequency for transmission over a wireless channel.Thus, it should be understood that front end 12 may be any component orcombination of components that produces a differential modulated inputsignal as broadly described in this disclosure.

In particular, when implemented with a continuous time switchedcapacitor network coupled to a physiological sensor, the continuous timeswitched capacitor network operates as a modulator that modulates(chops) the differential signal output by the physiological sensor to acarrier frequency. The physiological sensor may be a set of electrodes,an accelerometer, a pressure sensor, a voltage sensor or other sensorthat outputs a differential voltage signal. In particular, thephysiological sensor may, for example, generate a differential signalproportional to physiological signals such as, ECG signals, EMG signals,EEG signals, or other signals. The differential signal generated by thesensor is a low frequency signal. Using physiological signals as anexample, the frequency of the differential signal may be within a rangeof approximately 0 Hz to approximately 100 Hz, and may be less thanapproximately 2 Hz, and in some cases less than approximately 1 Hz.

Sensors other than physiological sensors may also be used. That is, thesensor does not need to output a differential signal proportional to aphysiological signal. Rather, the sensor may be any electrode,accelerometer, pressure sensor, voltage sensor or other sensor thatoutputs a differential signal, which may or may not represent aphysiological signal or serve a medical sensing application. However, inthe case of a physiological sensor, the carrier frequency may be withina range of approximately 4 kHz to approximately 10 kHz, although otherfrequencies are possible. It is important, however, that the carrierfrequency be sufficiently higher than the frequency of the basebandsignal of interest and within a range that does not introducesignificant noise into the signal, i.e., a frequency at which mixeramplifier 14 operates without introducing noise into the signal.

In this case, the modulator in front end 12 may include a differentialset of switches, e.g., CMOS switches, that are toggled between theoutputs of the physiological sensor to modulate (chop) an amplitude ofthe input signal. Clock 96 supplies the clock signal that the modulatorin the front end 12 and demodulator 86 in mixer amplifier 14 use tomodulate the differential input signal at the carrier (chop) frequency.At one end, the switches are cross coupled to each other and togglebetween the output terminals of the sensor to reject common mode signalsand operate as continuous time process, i.e., a non-sampling process.The switches are coupled at the other end to input capacitors of mixeramplifier 14 to form a continuous time switched capacitor network. Inthis way, front end 12 amplitude modulates (chops) the differentialinput signal at the inputs to mixer amplifier 14. Consequently, themodulated differential input signal produced by front end 12 is a squarewave with a frequency equal to the carrier frequency. A circuit diagramfor this example embodiment is provided in FIG. 8.

When front end 12 is implemented as an impedance sensor, front end 12may include a set of CMOS SPDT switches that are coupled at one end toreference potentials and to corresponding resistors at the other end.The switches toggle between the reference potentials and arecross-coupled to each other to modulate (chop) a stimulation currentthrough the resistors and reject common-mode signals. The resistors maybe connected in series to respective capacitors that are AC coupled tomixer amplifier 14 through tissue of a patient. The chopped stimulationcurrent produces a chopped voltage on the tissue with an amplitudemodulated at the carrier frequency that is AC coupled to mixer amplifier14. A circuit diagram is provided for this example embodiment in FIG. 9.

When instrumentation amplifier 10 is used to demodulate telemetrysignals, front end 12 may be viewed as part of a transmitter in thetelemetry system. In particular, front end 12 may be implemented usingany circuitry known in the art of telemetry that modulates a dataencoded signal to a carrier frequency for transmission over a wirelesschannel. For example, front end 12 may be viewed as part of a receiverlocated in an IPG that is implanted within a patient and communicateswith a clinician or patient programmer. Alternatively, front end 12 maybe part of a receiver of the clinician or patient programmer thatcommunicates with the IPG implanted within the patient. A detailed blockdiagram for this example embodiment is provided in FIG. 15A.

In any case, front end 12 generates a differential input signal formixer amplifier 14. Noise, e.g., 1/f noise, popcorn noise, and offset,enters the signal path of instrumentation amplifier 10 at mixeramplifier 14 to produce noisy modulated input signal 25. Noisy modulatedinput signal 25 includes the original low frequency components modulatedup to the carrier frequency and noise components at baseband.

As previously described, mixer amplifier 14 may be implemented using themodified folded-cascode amplifier architecture illustrated in FIG. 6.Reference and bias generator 94 supplies bias and reference voltages tomixer amplifier 14. In the interest of simplicity, mixer amplifier 14 isillustrated in FIG. 7 as including amplifier 84, demodulator 86, andintegrator 88, which correspond to amplifier 26, demodulator 28, andintegrator 30 in FIG. 2. Accordingly, amplifier 84 amplifies noisymodulated input signal 25 and demodulator 86 demodulates amplifiedsignal 27. More specifically, demodulator 86 demodulates the originallow frequency signal components of the amplified signal back down tobaseband and modulates noise 23 up to the carrier frequency, therebymaintaining separation between the desired signal and noise. Clock 96supplies a clock signal to drive demodulator 86. For example, withrespect to the circuit diagram of FIG. 6, clock 96 supplies a clocksignal to drive switches 60 and 62 which operate as demodulator 86.Integrator 88 integrates demodulated signal 29 with respect to areference voltage supplied by reference and bias generator 94 and actsas a low pass filter that substantially eliminates signal componentswith a frequency outside of the baseband. Consequently, noise sitting atthe carrier frequency of demodulated signal 29 is substantiallyeliminated from the output of integrator 88, i.e., output signal 31.

In FIG. 7, feedback 16 includes negative feedback path 90, negativefeedback path 92, and positive feedback path 98. To provide adifferential-to-single conversion, each of feedback paths 90, 92, and 98may include two symmetrical feedback path branches to provide feedbackto respective positive and negative differential inputs of mixeramplifier 14. In particular, negative feedback path 90 provides negativefeedback at the input to mixer amplifier 14 to keep the signal changesmall. Each of the feedback path branches of negative feedback path 90modulates output signal 31 with a reference voltage provided byreference and bias generator 94. To ensure that a negative feedback pathexists in negative feedback path 90 at all times, the chop frequencyapplied to the negative feedback path branches of feedback path 90should be 180 degrees out of phase with each other with one of thefeedback paths synchronous with front end 12. In this way, one of thefeedback path branches of negative feedback path 90 is applying negativefeedback during each half of the clock cycle. As a result, thedifferential signals at the input of mixer amplifier 14 are small andcentered about the reference voltage. Negative feedback 90 substantiallyeliminates the dynamic limitation of instrumentation amplifier 10, i.e.,glitching in output signal 31.

Negative feedback path 92 allows for the construction of a high passfilter. In particular, negative feedback path 92 integrates the outputof instrumentation amplifier 10, i.e., output signal 31, with respect toa reference voltage supplied by reference and bias generator 94 andapplies the integrated signal to the inputs of mixer amplifier 14through a capacitor. Each of the feedback path branches of negativefeedback path 92 modulates the integrated output signal with thereference voltage. Similar to the previously described feedback paths ofnegative feedback path 90, relative phasing of feedback path branches ofnegative feedback path 92 should ensure that a negative feedback pathexists for each half of the clock cycle. In operation, negative feedbackpath 92 is dominant at low frequency and suppresses the DC response ofinstrumentation amplifier 10. However, negative feedback path 90 isdominant at passband frequencies. The scaling of feedback capacitors infeedback path 90 and the time constant of feedback path 92 set the highpass corner of the filter. In other words, capacitors in feedback paths90 and 92 are used to set the high pass corner.

As an example, a high pass filter may be useful for rejectingpost-pacing artifacts when instrumentation amplifier 10 is used forheart monitoring applications and filtering out electrode offsets wheninstrumentation amplifier is used for monitoring brain signals. As anexample, feedback path 92 may be used to construct a high pass filterwith a cutoff frequency equal to approximately 2.5 Hz, 0.5 Hz, or 0.05Hz. In this case, feedback path 92 may be dominant at frequencies belowcutoff frequencies of 2.5 Hz, 0.5 Hz, or 0.05 Hz, while feedback path 90may be dominant at frequencies above the cutoff frequencies. In oneexample, feedback path 92 may have a cutoff frequency of approximately0.5 Hz, permitting feedback path 90 to dominate at frequencies aboveapproximately 0.5 Hz, e.g., approximately 5 Hz to 100 Hz

Positive feedback path 98 increases the input impedance ofinstrumentation amplifier 10. More specifically, positive feedback path98 samples output signal 31 and provides feedback to front end 12 beforechopper modulation is applied to the input signal. The positive feedbackeffectively “replaces charge” on the input capacitors to mixer amplifier14 that is lost during the sampling process. Positive feedback path 98may increase the input impedance of instrumentation amplifier 10 by anorder of magnitude or more. Each feedback path branch of positivefeedback path 98 may include a switched capacitor arrangement to addcompensatory charge to the input capacitors.

Although FIG. 7 depicts feedback path 16 as including negative feedbackpath 90, negative feedback path 92, and positive feedback path 98, onlynegative feedback path 90 may be provided for instrumentation amplifier10 to achieve stable measurements at low frequency with very low power.Accordingly, feedback paths 92, 98 may be considered optional, auxiliaryfeedback paths that enable instrumentation amplifier 10 to achieveadditional performance enhancements. Consequently, various exampleembodiments of the invention described in this disclosure may includeone, both, or neither of feedback paths 92, 98. When the instrumentationamplifier includes feedback paths 92 and 98, positive feedback path 98may sample the integrated output signal from negative feedback path 92instead of sampling the output signal of mixer amplifier 14. Therelative arrangement of feedback paths 90, 92, 98 may be more apparentfrom the circuit diagrams that follow in the additional figures.

In some embodiments, clock 96 may comprise one or more clocks. Forexample, when instrumentation amplifier 10 is implanted on a singlechip, a single clock may supply clock signals to front end 12, mixeramplifier 14, and feedback path 16. However, in some embodiments, suchas when instrumentation amplifier 10 is used to demodulate telemetrysignals, front end 12 may be implemented on a separate chip than mixeramplifier 14 and feedback 16. In this case, front end 12 may be suppliedwith a clock signal from one clock while a different clock providesclock signals to mixer amplifier 14 and feedback 16. In this case, thetwo clocks may not be in phase with each other. Since the clocks shouldbe in phase with each other to ensure that the transmitted signal can berecovered, additional circuitry may be required at the receiver tosynchronize the clocks.

Reference and bias generator 94 supplies bias voltages to front end 12,mixer amplifier 14, negative feedback path 90, and negative feedbackpath 92. When front end 12 includes a physiological sensor, referenceand bias generator 94 may supply reference voltages that drive thephysiological sensor. Reference and bias generator 94 may also supplythe reference voltages to electrodes for an impedance sensor. Withrespect to mixer amplifier 14, reference and bias generator 94 maysupply bias voltages for biasing the transistors as shown in FIG. 6. Thereference voltages that are mixed with the signals in feedback paths 90and 92 as previously described may also be supplied by reference andbias generator 94. Bias voltages of 0 volts to 1.2 volts (bandgap) or 0volts to 0.6 volts (half bandgap) may be used as bias points.

FIG. 8 is a circuit diagram illustrating an instrumentation amplifier100. Instrumentation amplifier 100 is an example embodiment ofinstrumentation amplifier 10 previously described in this disclosure. InFIG. 8, instrumentation amplifier 100 includes sensor 101 whichgenerates a differential voltage across its outputs 102A and 102B(collectively referred to as “outputs 102”). Outputs 102A and 102Bprovide voltages Vin-plus and Vin-minus, respectively. Sensor 101 may bea physiological sensor that translates biophysical signals to adifferential electrical voltage across outputs 102. For example, sensor101 may be an accelerometer, a pressure sensor, a force sensor, agyroscope, a humidity sensor, a pair of electrodes, or the like.

Inputs 102A and 102B are connected to capacitors 106A and 106B(collectively referred to as “capacitors 106”) through switches 104A and104B (collectively referred to as “switches 104), respectively. Switches104 are driven by a clock signal provided by a system clock (not shown)and are cross-coupled to each other to reject common-mode signals.Capacitors 106 are coupled at one end to a corresponding one of switches104 and to a corresponding input of mixer amplifier 116 at the otherend. In particular, capacitor 106A is coupled to the positive input ofmixer amplifier 116, and capacitor 106B is coupled to the negative inputof amplifier 116, providing a differential input.

In FIG. 8, sensor 101, switches 104, and capacitors 106 form front end110. Front end 110 generally corresponds to front end 12 ofinstrumentation amplifier 10. In particular, front end 110 operates as acontinuous time switched capacitor network as previously described withrespect to front end 12. Switches 104 toggle between an open state and aclosed state in which inputs 102 are coupled to capacitors 106 at aclock frequency to modulate (chop) the output of sensor 101 to thecarrier (clock) frequency. As previously described, the output of sensor101 may be a low frequency signal within a range of approximately 0 Hzto approximately 100 Hz. The carrier frequency may be within a range ofapproximately 4 kHz to approximately 10 kHz. Hence, the low frequencysensor output is chopped to the higher chop frequency band.

Switches 104 toggle in-phase with one another to provide a differentialinput signal to mixer amplifier 116. During a first phase of the clocksignal, switch 104A connects sensor output 102B to capacitor 106A andswitch 104B connects sensor output 102A to capacitor 106B. During asecond phase, switches 104 change state such that switch 104A couplesport 102A to capacitor 106A and switch 104B couples port 102B tocapacitor 106B. Switches 104 synchronously alternate between the firstand second phases to modulate the differential voltage at outputs 102 atthe carrier frequency. The resulting chopped differential signal isapplied across capacitors 106, which couple the differential signalacross the inputs of mixer amplifier 116.

Resistors 108A and 108B (collectively referred to as “resistors 108”)provide a DC conduction path that controls the voltage bias at the inputof mixer amplifier 116. In other words, resistors 108 may be selected toprovide an equivalent resistance that is used to keep the bias impedancehigh. Resistors 108 may, for example, be selected to provide a 5 GΩequivalent resistor, but the absolute size of the equivalent resistor isnot critical to the performance of instrumentation amplifier 100. Ingeneral, increasing the impedance improves the noise performance andrejection of harmonics, but extends the recovery time from an overload.To provide a frame of reference, a 5 GΩ equivalent resistor results in areferred-to-input (RTI) noise of approximately 20 nV/rt Hz with an inputcapacitance (Cin) of approximately 25 pF. In light of this, a strongermotivation for keeping the impedance high is the rejection of highfrequency harmonics which can alias into the signal chain due tosettling at the input nodes of mixer amplifier 116 during each half of aclock cycle.

It is important to note that resistors 108 are merely exemplary andserve to illustrate one of many different biasing schemes forcontrolling the signal input to mixer amplifier 116. In fact, thebiasing scheme is flexible because the absolute value of the resultingequivalent resistance is not critical. In general, the time constant ofresistor 108 and input capacitor 106 may be selected to be approximately100 times longer than the reciprocal of the chopping frequency.

Mixer amplifier 116 may produce noise and offset in the differentialsignal applied to its inputs. For this reason, the differential inputsignal is chopped via switches 104A, 104B and capacitors 106A, 106B toplace the signal of interest in a different frequency band from thenoise and offset. Then, mixer amplifier 116 chops the amplified signal asecond time to demodulate the signal of interest down to baseband whilemodulating the noise and offset up to the chop frequency band. In thismanner, instrumentation amplifier 100 maintains substantial separationbetween the noise and offset and the signal of interest. Mixer amplifier116 and feedback path 118 process the noisy modulated input signal toachieve a stable measurement of the low frequency signal output bysensor 101 while operating at low power.

As previously described, operating at low power tends to limit thebandwidth of mixer amplifier 116 and creates distortion (ripple) in theoutput signal. Mixer amplifier 116 and feedback path 118 correspond toand, thus, operate in a manner similar to previously described mixeramplifier 14 and feedback path 16. More specifically, feedback path 118corresponds to negative feedback path 90 described in FIG. 7 Mixeramplifier 116 and feedback path 118 substantially eliminate the dynamiclimitations of chopper stabilization through a combination of choppingat low-impedance nodes and AC feedback, respectively.

In FIG. 8, mixer amplifier 116 is represented with the circuit symbolfor an amplifier in the interest of simplicity. However, it should beunderstood that mixer amplifier 116 may be implemented in accordancewith the circuit diagram provided in FIG. 6. Consequently, mixeramplifier 116 provides synchronous demodulation with respect to frontend 12 and substantially eliminates 1/f noise, popcorn noise, and offsetfrom the signal to output a signal that is an amplified representationof the differential voltage produced by sensor 101.

Without the negative feedback provided by feedback path 118, the outputof mixer amplifier 116 would include spikes superimposed on the desiredsignal because of the limited bandwidth of the amplifier at low power.However, the negative feedback provided by feedback path 118 suppressesthese spikes so that the output of instrumentation amplifier 100 insteady state is an amplified representation of the differential voltageproduced by sensor 101 with very little noise.

Feedback path 118 in FIG. 8 may include two feedback paths that providea differential-to-single ended interface. The top feedback path branchmodulates the output of mixer amplifier 116 to provide negative feedbackto the positive input terminal of mixer amplifier 116. The feedback pathbranch includes capacitor 112A and switch 114A. Similarly, the bottomfeedback path branch of feedback path 118 includes capacitor 112B andswitch 114B that modulate the output of mixer amplifier 116 to providenegative feedback to the negative input terminal of mixer amplifier 116.Capacitors 112A and 112B are connected at one end to switches 114A and114B, and at the other end to the positive and negative input terminalsof mixer amplifier 116, respectively.

Switches 114A and 114B toggle between a reference voltage (Vref) and theoutput of mixer amplifier 116 to place a charge on capacitors 112A and112B, respectively. The reference voltage may be, for example, amid-rail voltage between a maximum rail voltage of amplifier 116 andground. For example, if the amplifier circuit is powered with a sourceof 0 to 2 volts, then the mid-rail Vref voltage may be on the order of 1volt. Importantly, switches 114A and 114B should be 180 degrees out ofphase with each other to ensure that a negative feedback path existsduring each half of the clock cycle. One of switches 114 should also besynchronized with mixer amplifier 116 so that the negative feedbacksuppresses the amplitude of the input signal to mixer amplifier 116 tokeep the signal change small in steady state. By keeping the signalchange small and switching at low impedance nodes of mixer amplifier116, e.g., as shown in the circuit diagram of FIG. 6, the onlysignificant voltage transitions occur at switching nodes. Consequently,glitching (ripples) is substantially eliminated or reduced at the outputof mixer amplifier 116.

Switches 104 and 114, as well as the switches at low impedance nodes ofmixer amplifier 116, may be CMOS SPDT switches. CMOS switches providefast switching dynamics that enables switching to be viewed as acontinuous process. The transfer function of instrumentation amplifier100 may be defined by the transfer function provided in equation (1)below, where Vout is the voltage of the output of mixer amplifier 116,Cin is the capacitance of input capacitors 106, ΔVin is the differentialvoltage at the inputs to mixer amplifier 116, Cfb is the capacitance offeedback capacitors 112, and Vref is the reference voltage that switches114 mix with the output of mixer amplifier 116.

Vout=Cin(ΔVin)/Cfb+Vref  (1)

From equation (1), it is clear that the gain of instrumentationamplifier 100 is set by the ratio of input capacitors Cin and feedbackcapacitors Cfb, i.e., capacitors 106 and capacitors 112. The ratio ofCin/Cfb may be selected to be on the order of 100. Capacitors 112 may bepoly-poly, on-chip capacitors or other types of MOS capacitors andshould be well matched, i.e., symmetrical.

Although not shown in FIG. 8, instrumentation amplifier 100 may includeshunt feedback paths for auto-zeroing amplifier 100. The shunt feedbackpaths may be used to quickly reset amplifier 100. An emergency rechargeswitch also may be provided to shunt the biasing node to help reset theamplifier quickly. The function of input capacitors 106 is toup-modulate the low-frequency differential voltage from sensor 101 andreject common-mode signals. As discussed above, to achieveup-modulation, the differential inputs are connected to sensingcapacitors 106A, 106B through SPDT switches 104. The phasing of theswitches provides for a differential input to the ac transconductancemixing amplifier 116. These switches 104 operate at the clock frequency,e.g., 4 kHz. Because the sensing capacitors 106 toggle between the twoinputs, the differential voltage is up-modulated to the carrierfrequency while the low-frequency common-mode signals are suppressed bya zero in the charge transfer function. The rejection ofhigher-bandwidth common signals relies on this differential architectureand good matching of the capacitors.

As further shown in FIG. 8, for applications in which measurements aretaken in conjunction with stimulation pulses delivered by a cardiacpacemaker, cardiac defibrillator, or neurostimulator, blanking circuitrymay be added to instrumentation amplifier 100 the inputs of mixeramplifier 116 and coupling capacitors 106 to ensure that the inputsignal settles before reconnecting mixer amplifier 116 to front end 110.For example, the blanking circuitry may be a blanking multiplexer (MUX)111 that selectively couples and de-couples mixer amplifier 116 fromfront end 110. This blanking circuitry selectively decouples the mixeramplifier from the differential input signal and selectively disablesthe first and second modulators, i.e., switches 104, 114, e.g., duringdelivery of a stimulation pulse.

Blanking MUX 111 is optional but may be desirable. The clocks drivingswitches 104, 114 to function as modulators cannot be simply shut offbecause the residual offset voltage on mixer amplifier 116 wouldsaturate the amplifier in a few milliseconds. For this reason, blankingMUX 111 may be provided to decouple amplifier 116 from the input signalfor a specified period of time during and following application of astimulation by a cardiac pacemaker or defibrillator, or by aneurostimulator.

To achieve suitable blanking, the input and feedback switches 104, 114should be disabled while mixer amplifier 116 continues to demodulate theinput signal. This holds the state of the integrator within mixeramplifier 116 because the modulated signal is not present at the inputsof the integrator, while the demodulator continues to chop the DCoffsets. Accordingly, blanking MUX 111 may further include circuitry orbe associated with circuitry configured to selectively disable switches104, 114 during a blanking interval. Post blanking, mixer amplifier 116may require additional time to resettle because some perturbations mayremain. Thus, the total blanking time includes time for demodulating theinput signal while the input and switches 104, 114 are disabled and timefor settling of any remaining perturbations. An example blanking timefollowing application of a stimulation pulse may be approximately 8 mswith 5 ms for mixer amplifier 116 and 3 ms for the AC couplingcomponents.

FIG. 9 is a circuit diagram illustrating an instrumentation amplifier200 for measuring impedance across a tissue load 211. Tissue load 211represents the tissue of a patient for which impedance is measured byinstrumentation amplifier 200. Tissue 211 may be organ tissue, such asheart tissue, lung tissue, or brain tissue, muscle tissue, adiposetissue, or other tissue for which the impedance may be measured todiagnose chronic or acute disease states or other medical conditions.Some example applications for impedance measurements include detectionof pulmonary edema, minute ventilation measurements for respiration,measurement of cardiac dynamics, and measurement of brain signals. Ingeneral, it is important that instrumentation amplifier 200 does notstimulate excitable cells in the tissue or cause other detrimentaleffects such as electrode corrosion.

Instrumentation amplifier 200 may generally conform to instrumentationamplifier 10 described with reference to FIGS. 1-7. In the example ofFIG. 9, instrumentation amplifier 200 applies synchronous detectionprinciples to accurately measure the impedance of tissue load 211 withlow power, inherent charge balancing, rejection of electrode potentials,and small stimulation currents. Instrumentation amplifier 200 is anexample embodiment of previously described instrumentation amplifier 10.Like instrumentation amplifier 10, instrumentation amplifier 200includes a front end 210, mixer amplifier 226, and feedback path 228.These features may generally correspond to front end 12, mixer amplifier14, and feedback path 16 of instrumentation amplifier 10.

In FIG. 9, front end 210 includes input voltages at ports 202A and 202B(collectively referred to as “ports 202”), switches 204A and 204B(collectively referred to as “switches 204”), resistors 206A and 206B(collectively referred to as “resistors 206”), and capacitors 208A and208B (collectively referred to as “capacitors 208”). In general, frontend 210 modulates a stimulation current that creates a voltage on tissueload 211. The stimulation current may be applied across tissue load 211via two or more electrodes, which may be mounted on one or more leads orcarried on a surface of an implantable medical device housing.Similarly, the resulting voltage signal across tissue load 211 may besensed by two or more electrodes deployed on one or more leads or on adevice housing. The voltage on tissue load 211 is AC coupled to positiveand negative inputs of mixer amplifier 226 by capacitors 222A and 222B(collectively referred to as “capacitors 222”), respectively. Thus, thetissue represented by tissue load 211 is not exposed to DC current.Moreover, the small modulated (AC) stimulation current, which may beapproximately 10 μA or less, may not substantially excite the tissuerepresented by tissue load 211.

Switches 204 toggle between input voltages at ports 202 (Vstim+ andVstim−) to generate stimulation current through resistor-capacitor (RC)pairs of resistor 206A and capacitor 208A and resistor 206B andcapacitor 208B. Switches 204, resistors 206 and capacitors 208 may forman alternating current (ac) source that generates an ac stimulationcurrent at a clock frequency for application to a load, such as 211. Inparticular, switches 204, resistors 206 and capacitors 208 form amodulator that modulates first and second voltages Vstim+ and Vstim− atthe clock frequency to produce the stimulation current for applicationto the load. However, other types of ac current sources may be use toprovide the ac stimulation current for impedance measurement.

The input voltages Vstim+ and Vstim− may be provided by regulated powersupplies within a device in which instrumentation amplifier 200 isemployed, such as an implantable medical device. Switches 204 open andclose at a chopper frequency to, in effect, chop the input stimulationcurrent delivered by input voltages at ports 202 via the RC pairs (206,208) to measure tissue impedance. In this manner, front end 210generates a modulated differential input signal that is processed bymixer amplifier 226 and feedback path 228. Stimulation currents at ports202 may be provided by electrodes carried on leads that are connected toan IPG implanted within a patient. This is one example of delivery ofstimulation current for impedance measurements. As an alternative,stimulation current for impedance measurement could be generated by oneor more switched current sources. The reference voltages at ports 202and the sizes of resistors 206 and capacitors 208 may be determined bythe constraints on the stimulation current, linearity of themeasurement, and the time constant of instrumentation amplifier 200compared to the clock (not shown) that drives switches 204.

As an example, using a stimulation current of 10 μA, voltages at ports202A and 202B may provide 2V and 0 V, respectively, and resistors 206may be selected as 100 kΩ resistors. Alternatively, using 2000 kΩresistors yields a 0.5 μA stimulation current with 100 kΩ resistors.Using 10 nF capacitors for capacitors 208 results in a stimulationcurrent having a time constant of 1 ms, which requires a stimulationcurrent with a frequency of approximately 5 kHz to ensure minimal errorfrom settling dynamics. The nonlinearity of the measurement, assuming 1kHz loads, is bounded to under 0.5% in this case.

The input to mixer amplifier 226 may include a high pass filter 212 andcoupling capacitors 222A, 222B. In some embodiments, high pass filter212 assists in keeping post-pace recovery to a minimum for cardiacdynamic measurements. In FIG. 9, high pass filter 212 includescapacitors 214A, 214B (collectively referred to as “capacitors 214”) andresistors 216A, 216B (collectively referred to as “resistors 216”). Thevalues of capacitors 214 and resistors 216 may be selected such thathigh pass filter 212 has a high pass corner frequency that ensuresminimal phase error, e.g., less than 1% equivalent measurement error,occurs at mixer amplifier 226 while settling any residual pacing errorsin 2.5 ms to 5 time constants. For some applications, such as cardiacimpedance analysis, the high pass corner frequency may, for example, bewithin a range of approximately 300 Hz to approximately 800 Hz.

Resistors 224A and 224B (collectively referred to as “resistors 224”)control the voltage at the input of mixer amplifier 226. Accordingly,resistors 224 are similar to resistors 108 in FIG. 7 and are merelyexemplary. As previously described, resistors 224 or a different biasscheme may be selected to provide a 5 GΩ equivalent resistor althoughthe absolute value is not critical.

Mixer amplifier 226 and feedback path 228 process the noisy modulatedinput signal to achieve a stable measurement of the differential voltageon tissue load 211 while operating at low power. Mixer amplifier 226 andfeedback path 228 generally correspond to mixer amplifier 116 andfeedback path 118 in FIG. 7. Accordingly, mixer amplifier 226 providessynchronous demodulation with respect to front end 12 and substantiallyeliminates noise, i.e., 1/f noise, popcorn noise, and offset, from theamplified output signal. Mixer amplifier 226 may be implemented usingthe modified folded-cascode architecture with switching at low impedancenodes, e.g., substantially as shown in FIG. 6.

As shown in FIG. 9, feedback path 228 includes top and bottom feedbackpath branches that provide negative feedback and asingle-to-differential interface. The top and bottom feedback pathbranches include capacitors 230A and 230B (collectively referred to as“capacitors 230”) which are connected to switches 232A and 232B(collectively referred to as “switches 232”), respectively. Switches232A and 232B are 180 degrees out of phase with each other and togglebetween the output of mixer amplifier 226 and a reference voltage (Vref)to modulate the output of mixer amplifier 226. Consequently, feedbackpath 218 provides negative feedback to keep the signal change at theinput to mixer amplifier 226 small as previously described in thisdisclosure.

Switches 206, switches 232, and the switches at low impedance nodes inmixer amplifier 226 may be CMOS SPDT switches or other switches thatprovide fast switching dynamics. The transfer function forinstrumentation amplifier 200 is the same as that for instrumentationamplifier 100, which is provided in the above description of FIGS. 7 and8. Thus, the ratio of the capacitance of feedback capacitors, i.e.,capacitors 230, to the capacitance of input capacitors, i.e., capacitors222, sets the gain of instrumentation amplifier 226. Capacitors 222 and230 may be poly-poly capacitors or other types of MOS capacitors andshould be well matched, i.e., symmetrical. Capacitors 222 and 230 may beplaced on chip with the other instrumentation amplifier components.

In operation, instrumentation amplifier 200 may fold electromagneticinterference (EMI) into the modulated input signal at the carrierfrequency and odd harmonics. In order to determine if the channel iscorrupt, the output of instrumentation amplifier 200 can be monitoredwith no stimulation current applied to front end 210. Alternatively,spread-spectrum techniques may be used to break up the synchronous clockdetection between front end 210 and mixer amplifier 226. Spread-spectrumclocking breaks up the uncorrelated noise into a broadband noise signalthat is substantially eliminated by mixer amplifier 226, whilemaintaining the correlated impedance measurement.

The output of instrumentation amplifier 200 may be sent to ananalog-to-digital converter (ADC) (not shown) that applies additionalprocessing for measuring the impedance of tissue load 211. Further, wheninstrumentation amplifier 200 is implanted within a patient, thetissue-electrode interface (front end 12) may be galvanically isolatedfrom the measurement circuit (mixer amplifier 226 and feedback path228). Isolation helps to reject electrode polarization and ensure netcharge balance across the electrodes.

Instrumentation amplifier 200 can be used to separate the measurement oflead impedances from impedance measurements for edema, minuteventilation, and cardiac dynamics. The reason for this is that therequirements are different for the two measurements. Lead impedancesordinarily require a quick sample to be taken just prior to delivery ofa pacing or stimulation pulse, with several vectors requiringmeasurement. Perturbation of the sensing channel is not a major issuesince the stimulation pulse immediately follows the measurement. Thisfavors the application of large, fast, sampled stimulation current. Themeasurement of edema, minute ventilation and cardiac dynamics, however,occur at low frequency where the sensing channel should be free ofperturbations and noise. Significant perturbations from thismeasurement, i.e., measurement of lead impedances, compromises theability of the sense channel to accurately detect evoked potentialspost-pace and can result in oversensing. Edema, minute ventilation andcardiac dynamic measurements therefore favor low level stimuli, averagedwith continuous time methods. Instrumentation amplifier 200 enablesedema, minute ventilation and cardiac dynamic measurements to beseparate from measurement of lead impedances.

Although not shown in FIG. 9, for applications in which measurements aretaken in conjunction with stimulation pulses delivered by a cardiacpacemaker or neurostimulator, blanking circuitry such as the blankingMUX 111 shown in FIG. 8 may be added to instrumentation amplifier 200.For example, a blanking MUX may disconnect input capacitors 222 from theinputs of mixer amplifier 226. In addition, input and feedbackmodulators may be disabled during the blanking period. In someembodiments, the blanking MUX may be placed between high pass filter 212and coupling capacitors 222 to ensure that the input signal settlesbefore reconnecting mixer amplifier 226 to front end 210. Hence, theblanking circuitry may be a multiplexer (MUX) that selectively couplesand de-couples mixer amplifier 226 from front end 210. As mentioned withreference to FIG. 8, blanking circuitry may be desirable because theclocks driving the switches cannot be simply shut off since the residualoffset voltage on mixer amplifier 226 would saturate the amplifier in afew milliseconds.

To achieve suitable blanking, the input and feedback switches 222, 232,should be disabled while mixer amplifier 226 continues to demodulate theinput signal. This holds the state of the integrator within mixeramplifier 226 because the modulated signal is not present at the inputsof the integrator, while the demodulator continues to chop the DCoffsets. Post blanking, mixer amplifier 226 may require additional timeto resettle because some perturbations may remain. Thus, the totalblanking time includes time for demodulating the input signal while theinput and feedback switches are disabled and time for settling of anyremaining perturbations. An example blanking time may be approximately 8ms with 5 ms for mixer amplifier 226 and 3 ms for the AC couplingcomponents.

Through experimentation, it has been found that the linearity ofmeasurement via instrumentation amplifier 200 meets a theoretical limitof 0.05% for a 500 nA stimulation current and 1.5% for a 10 μAstimulation current. The worst-case linearity is at high impedance, dueto finite output impedance of mixer amplifier 226. In other words,higher stimulation currents result in greater non-linearity. Inpractice, the observable nonlinearity is small for reasonablestimulation vectors through a tissue load on the order of 1 kΩ.

Experimentation has also shown the measured noise floor of aninstrumentation amplifier including a mixer amplifier and negativefeedback, such as instrumentation amplifier 100 and 200, to beapproximately 100 nV/rt Hz. This is in line with theoreticalexpectations form Johnson noise in the input transistors of the mixeramplifier 226 operating with 1 μA of stimulation current. For a 10 μAstimulation current, this translates into an equivalent noise floor of0.01 ohms/rtHz, which is well below the requirements in manyphysiological applications.

FIG. 10 is a diagram illustrating an example signal flow for aninstrumentation amplifier 300 that includes negative feedback forconstructing a high pass filter. With respect to FIG. 2, thearchitecture of instrumentation amplifier 300 in FIG. 10 may besubstantially the same as that of instrumentation amplifier 10, but withthe addition of negative feedback path 92. Accordingly, similar numbercomponents in FIG. 2 and FIG. 10 share similar functionality. In theinterest of brevity and to avoid redundancy, the signal flow throughfront end 10, mixer amplifier 14 and feedback path 90 is not describedin detail. Instead, the flow of output signal 31 which is produced bymixer amplifier 14 through negative feedback path 92 is described.

In general, negative feedback path 92 performs additional signalprocessing on output signal 31 to construct a high pass filter at theinput to mixer amplifier 14. The high pass filter substantiallyeliminates signal components that have a frequency below the cornerfrequency of the high pass filter. For example, feedback path 92 may seta corner frequency of approximately equal to 2.5 Hz, 0.5 Hz, or 0.05 Hz.In general, negative feedback path 92 suppresses signals between thecorner frequency and DC. As previously described, feedback path 92provides differential feedback to respective input terminals of mixeramplifier 14 through symmetrical feedback paths. The feedback pathsshould be 180 degrees out of phase with each other so that negativefeedback is applied during each half cycle of the clock cycle.

As shown in FIG. 10, negative feedback path 92 includes an integrator302 and modulator 304. Integrator 302 integrates output signal 31 withrespect to a reference voltage. This reference voltage should be thesame reference voltage that is modulated with the signal ininstrumentation amplifier 300 by modulators 20, 28, and 34. In someembodiments, a switched capacitor integrator may be used for integrator302. In other embodiments, a standard RC integrator may be used. Theswitched capacitor integrator may, however, provide certain advantages.

Modulator 304 modulates the output of integrator 302 to provide adifferential voltage into mixer amplifier 14. Since modulator 304 shouldbe synchronized with feedback path 90, clock signal 21C also drivesmodulator 304. Clock signal 21C is also supplied to integrator 302, asshown in FIG. 10, when integrator 302 is implemented as a switchedcapacitor integrator. Also shown in FIG. 10 are input capacitance (Cin)13, feedback capacitance (Cfb) 17 for feedback path 90, high pass filtercapacitance (Chp) 10 for feedback path 92.

In operation, integrator 302 produces a voltage on the switchedcapacitor of modulator 304 that counters the charge on the switchedcapacitor of modulator 34. When an input step is applied to mixeramplifier 14, the signal is integrated by integrator 30. Initially, thevoltage difference between demodulated signal 29 and the referencevoltage of integrator 30 is relatively large. In contrast, thedifference between the voltage of output signal 31 and the referencevoltage for integrator 302 is relatively small. As a result, integrator30 builds up charge on the switched capacitor of modulator 34 morequickly than integrator 302 builds up charge on the switched capacitorof modulator 304.

Over time, however, the voltage difference between demodulated signal 29and the reference voltage at integrator 30 decreases and integrator doesnot build up as much charge. At the same time, the voltage differencebetween output signal 31 and the reference voltage at integrator 302increases and integrator 302 builds up more charge on the switchedcapacitor at modulator 304. Thus, in steady state, feedback path 92dominates feedback path 90 and the feedback counter charge is mostlyprovided via negative feedback path 92. As a result, feedback path 92can set the high pass corner through a ratio of capacitors 17 and 19(Cfb and Chp) and a time constant set by the capacitors and clockfrequency of integrator 302. Importantly, since instrumentationamplifier 300 may be implemented entirely on a single chip, off chipcapacitors may not be needed for high pass filtering.

FIG. 11 is a circuit diagram illustrating instrumentation amplifier 300.As shown in FIG. 11, the architecture of instrumentation amplifier 300is substantially the same as that of instrumentation amplifier 100, butwith the addition of negative feedback path 92. Accordingly, similarnumber components in FIG. 7 and FIG. 10 share the same functionality.The operation of these shared components is not described in theinterest of brevity and to avoid redundancy. However, the operation offeedback path 92 is described.

Negative feedback path 92 taps off of the output of mixer amplifier 116and applies negative feedback to the inputs of mixer amplifier 116. Inthe example of FIG. 11, integrator 302 is a switched capacitorintegrator. Integrator 302 may be in addition to the integrator anddemodulator provided within mixer amplifier 116. The switched capacitorintegrator includes a capacitor 310 coupled between the output ofamplifier 116 and ground via switch 312A, and between the negative inputof amplifier 316 and ground via switch 312B. Switch 312A and 312B toggleat the chop frequency, but are out of phase with one another. Theclocking frequency of switches 312A and 312B can be adjusted to set thetime constant of integrator 302. The positive terminal of amplifier 316is coupled to a reference voltage (Vref), which may be the samereference voltage that is mixed with the signal at other stages ininstrumentation amplifier 300. Capacitor 314 couples the output ofamplifier 316 to the negative terminal of amplifier 316.

The two feedback paths of feedback path 92 tap off of the output ofintegrator 302 to provide negative feedback to mixer amplifier 116. Inparticular, the top feedback path branch modulates the output ofintegrator 302 to provide negative feedback to the positive terminal ofmixer amplifier 116. The top feedback path branch includes capacitor320A and switch 322A. Similarly, the bottom feedback path branch offeedback path 92 includes capacitor 320B and switch 322B, which modulatethe output of integrator 302 to provide negative feedback to thenegative terminal of mixer amplifier 116.

Capacitors 320A and 320B are connected at one end to switches 322A and322B, and to the positive and negative input terminals of mixeramplifier 116 at the other end, respectively. Switches 322A and 322Btoggle between a reference voltage (Vref) and the output of mixerintegrator 302 to place a charge on capacitors 320A and 320B,respectively. Switch 322A and 322B toggle 180 degrees out of phase withone another. Importantly, switches 322A and 322B should be synchronizedwith switches 114A and 114B, respectively. In this way, a negativefeedback path exists during each half cycle of the clock signal and issynchronized with the negative feedback path.

As previously described in FIG. 10, integrator 302 builds up a voltagethat is placed on capacitors 320A and 320B (collectively referred to as“capacitors 320”) by switches 322A and 322B (collectively referred to as“switches 322”). The charge on capacitors 320 counters the charge oncapacitors 106 in steady state. More specifically, the charge oncapacitors 320 dominates the feedback path in steady state for lowfrequencies. Thus, current substantially flows through negative feedbackpath 92 at steady state and little or no current flows through negativefeedback path 118. As a result, the ratio of feedback capacitors 112 and320 and the time-constant of integrator 302 sets the corner frequency ofthe high pass filter provided by negative feedback path 92. The cornerfrequency may be set to equal to approximately 2.5 Hz, 0.5 Hz, or 0.05Hz, or other desired frequencies. With feedback capacitors 112 on chip,the high-pass filter characteristics can be dynamically changed to helpwith recovery from an overload or transient.

Switches 312 and 322 may be CMOS SPDT switches or other switches thatprovide fast switching dynamics. Capacitors 310, 314, and 320 may bepoly-poly capacitors or other types of MOS capacitors.

It should be understood that feedback path 92, as shown in FIG. 11, maybe generally applied to an instrumentation amplifier as broadlydescribed in this disclosure. Accordingly, instrumentation amplifier 300should not be considered limiting in any way. Instead, instrumentationamplifier 300 is one of many example instrumentation amplifiers that mayinclude a negative feedback path for constructing a high pass filter asdescribed in this disclosure. For example, feedback path 92, as shown inFIG. 11, may be added to instrumentation amplifier 200 in FIG. 9.

FIG. 12 is a diagram illustrating an exemplary signal flow for aninstrumentation amplifier 400 that includes a positive feedback path forincreasing the input impedance of the instrumentation amplifier. Thearchitecture of instrumentation amplifier 400 may be substantially thesame as that of instrumentation amplifier 10 with respect to FIG. 2, butwith positive feedback path 98 included to provide additional signalprocessing. Accordingly, similar numbered components in FIG. 12 sharethe same functionality of those in FIGS. 2 and 10. In the interest ofbrevity and to avoid redundancy, the signal flow through front end 10,mixer amplifier 14, and feedback path 90 is not described in detail.Instead, the flow of output signal 31 which is produced by mixeramplifier 14 through positive feedback path 98 is described.

In general, positive feedback path 98 taps off of the output of mixeramplifier 14 or optionally the output of the integrator 302 in feedbackpath 92, if provided. Positive feedback path 98 provides feedback tofront end 12 prior to modulator 20, i.e., prior to application ofchopping input signal 32. As shown in FIG. 12, positive feedback path 98includes a switched capacitor arrangement 404 (Cpos) that is driven byclock signal 21C. In particular, switched capacitor 404 is used tocreate a resistance that is substantially equal to the effectiveresistance at the input of instrumentation amplifier 400. The effectiveinput resistance (Reff) of instrumentation amplifier is given inequation (2) below, where the frequency of clock signals 21A-C isFclock, and Cin is the capacitance of the input capacitors 106A, 106B atmodulator 20. Accordingly, the charge draw looking into instrumentationamplifier 400 is described by equation (3), where Q is the electriccharge, and ΔVin is the change in voltage.

$\begin{matrix}{{Reff} = {1/\left( {{Fclock} \cdot {Cin}} \right)}} & (2) \\{\frac{Q}{f} = {{{Cin} \cdot {Fclock} \cdot \Delta}\; {Vin}}} & (3)\end{matrix}$

Positive feedback path 98 compensates for the current passing throughthe effective resistance by “replacing” or putting charge back onto theswitched input capacitors 13 of modulator 20. Because the output voltageof instrumentation amplifier 400 without feedback path 98 isproportional to the differential input voltage multiplied by the ratioof the capacitance Cin of the input capacitors 106A, 106B of modulator20 to the capacitance Cfb of the feedback capacitors 112A, 112B ofmodulator 34, switched capacitor arrangement 404 (Cpos) samples outputof mixer amplifier 14 and uses positive feedback to replace the lostcharge. In other words, positive feedback path 98 injects current thatcompensates for current passing through the effective input resistance.Positive feedback path 98 may raise the equivalent low frequency inputimpedance by an order of magnitude or more.

Positive feedback path 98 may also be used at the same time as positivefeedback path 92. In this case, positive feedback path 98 may tap off ofthe output of the integrated signal output by positive feedback path 92.With respect to FIG. 10, positive feedback path 98 could tap off of theoutput of integrator 302, rather than the output of mixer amplifier 116.

FIG. 13 is a circuit diagram illustrating instrumentation amplifier 400.In FIG. 13, the architecture of instrumentation amplifier 400 issubstantially identical to that of instrumentation amplifier 300, butwith positive feedback path 98 tapping off of the output of mixeramplifier 116 and providing positive feedback to capacitors 106 of frontend 110. Components that share number between FIG. 13 and FIGS. 8 and 11share the same functionality. Accordingly, the operation of thesecomponents is not described in the interest of brevity and to avoidredundancy. However, the operation of positive feedback path 98 isdescribed.

In FIG. 13, positive feedback path 98 provides differential feedbackthrough a first feedback path branch and a second feedback path branch.The first feedback path branch (top branch) modulates the output ofmixer amplifier 116 to provide positive feedback to the positive inputterminal of mixer amplifier 114. The first feedback path branch (topbranch in FIG. 13) includes capacitor 410A, switch 412A, and switch412B. Switch 412A selectively couples one side of capacitor 410A toeither a reference voltage Vref or the output of mixer amplifier 116.Switch 412B selectively couples the other side of capacitor 410A toeither Vref or input port 102A of sensor 101. The second feedback pathbranch (bottom branch in FIG. 13) includes capacitor 410B and switch412C. One side of capacitor 410B is coupled to ground. Switch 412Cselectively couples the other side of capacitor 410B to either theoutput of mixer amplifier 116 or input port 102B of sensor 101.

Capacitors 410A and 410B are both coupled to the output of mixeramplifier 116 during a first clock phase. Thus, during the first clockphase, capacitors 410A and 410B sample the output of mixer amplifier116. One end of capacitor 410A is coupled to Vref during the firstphase. During a second clock phase, capacitors 410A and 410B are coupledat one end to input ports 102A, 102B, respectively. At the other end,during the second clock phase, capacitor 410A is coupled to Vref, whilecapacitor 410B is coupled to ground. The sizes of capacitors 410A and410B are selected according to the charged needed to compensate for thesampling of the input capacitors 106A, 106B during front end modulation.As an example, each capacitor 410A, 410B may have a capacitance valuethat is approximately twice the value of the feedback capacitance Cfb ofeach respective feedback capacitor 112A, 112B. Capacitors 410A, 410B maybe provided on-chip for close matching to capacitors 106A, 106B and112A, 112B.

In the second feedback path branch (bottom), charge is delivered to thefront end switch 104 b during a second clock phase, i.e., after thefirst clock phase in which capacitor 410B is coupled to sample theoutput of mixer amplifier 116. Similarly, in the first feedback pathbranch (top), charge is delivered to front end switch 104A during thesecond clock phase. To create a differential charge transfer from thesingle ended output of mixer amplifier 116, a different switching schemeis employed in the first feedback path branch (top) than in the bottomfeedback path branch). The clock frequency used to actuate switches412A, 412B, 412C may be the same as the chopping frequency. Thereference voltages used for feedback path 98, and particularly thereference voltages to which capacitor 410A is coupled in phase 1 andphase 2, should match the reference voltage used in feedback path 118.

Switches 412A, 412B and 412C may be CMOS SPDT switches or other switchesthat provide fast switching dynamics. Capacitors 410A and 410B may bepoly-poly capacitors or other types of MOS capacitors, and may be formedon-chip with capacitors 112A, 112B, 106A and 106B.

As previously described, positive feedback path 98 may also be used withnegative feedback path 92 at the same time. In this case, using FIG. 11as a reference, positive feedback path 98 could sample off of the outputof integrator 302. That is, switches 412A and 412C could be connected tothe output of integrator 302 instead of the output of mixer amplifier116.

FIG. 14A is a diagram illustrating the signal flow for aninstrumentation amplifier 500 that is used as part of a receiver 498 ina telemetry system. Instrumentation amplifier 500 may be used, forexample, as part of a receiver 498 in an implantable pulse generator(IPG), implantable drug delivery device, or other type of implantablemedical device (IMD) implanted within a patient that communicates, viatelemetry, with an external programming device, such as a clinician orpatient programmer. In addition, instrumentation amplifier 500 may alsobe located in an external programming device that communicates with anIPG or other type of IMD implanted within the patient. Receiver 498 mayreceive signals from a transmitter 499 associated with an IMD orexternal programmer. Receiver 498 and transmitter 499 together form atelemetry system that makes use of an instrumentation amplifier 500 asdescribed in this disclosure. As will be described, a first chopperstage resides in the transmitter 499 while a second chopper stage andfeedback path reside in instrumentation amplifier 500 in receiver 498.

In general, instrumentation amplifier 500 may be implemented as part oftelemetry circuitry in an IMD or programming device for an IMD thatcommunicates using “arms length telemetry.” Arms length telemetry (ALT)refers to telemetry over distances of approximately 10 cm or greater.For example, ALT may operate over a distance of approximately 50 cm or adistance of approximately 1 meter. Accordingly, ALT eliminates theburden of placing a programming device directly over the IMD forcommunication. However, the signal level for ALT is on the order ofhundreds of microvolts as a result of the signal level dropping off as acubic power of distance between the programming device and the IMD.Consequently, ALT requires micropower circuits to extract thetransmitted signal while suppressing or rejecting out of bandaggressors, i.e., noise. Aggressors include stimulation loop aggressorsand similar phenomena.

Instrumentation amplifier 500 may be configured to provide synchronousdemodulation for the detection of on-off-keyed (OOK) signals. As anexample, such signals may be transmitted by transmitter 499 in a 175 kHzindustry-scientific-medical (ISM) band. The chopper stabilized mixeramplifier described in this disclosure, i.e., mixer amplifier 14 withnegative feedback path 90, can be implemented in instrumentationamplifier 500 to provide synchronous demodulation with very low offsetand stable gain. Moreover, the gain of instrumentation amplifier 500 canbe conveniently determined by on-chip capacitor ratios, i.e., the ratioof the capacitance of the feedback capacitors in negative feedback path90 to the capacitance of the input capacitors. As shown in FIG. 14A,instrumentation amplifier 500 also includes a clock synchronizer 502 tocorrect for phase mismatch between clocks at the transmitter 499 andreceiver 498. Clock synchronizer 502 may include another chopperstabilized mixer amplifier in accordance with an embodiment of thisdisclosure.

In one example embodiment, the received signals may be transmitted usingon-off keying of a 175 kHz signal to send data between a programmingdevice and the IMD in which a receiver 498 incorporating instrumentationamplifier 500 resides. The 175 kHz signal falls within the ISM band. Thedata may be framed with a fixed interval of 22 μs to provide a 4.4 kbpsrate. The duty cycle of the signal within the frame signifies whetherthe data bit is a one or a zero.

It should be understood that instrumentation amplifier 500 is notlimited to the above protocol. Instead, this protocol is one of manyexample protocols that may be used for ALT. Accordingly, instrumentationamplifier 500 and the signal flow for instrumentation 500 in FIG. 14Ashould be viewed as examples for broadly teaching how a chopperstabilized instrumentation amplifier 500 described in this disclosurecan be used for synchronous demodulation of signals for arms lengthtelemetry and, therefore, should not be considered limiting in any way.

The signal flow of instrumentation amplifier 500 in FIG. 14A begins withtransmitter 499, which includes modulator 520. Modulator 520 receives aninput data signal 532 containing data to be transmitted, and chops theinput signal at a chopping frequency defined by clock signal 521A toproduce an output signal for transmission to receiver 498 via transmitantenna 501 and receiver antenna 503. Additional amplifier or filtercomponents may be provided to permit transmission of the modulatedsignal produced by modulator 520 In terms of an analog to the otherinstrumentation amplifier embodiments described in this disclosure,transmitter 499 and modulator 520 form, in effect, a front end 12 thatprovides the first chopping stage for the signal flow. Hence, in thiscase, front end 12 of the overall instrumentation amplifier 500 is atransmitter 499 associated with a separate device, e.g., an IMD orprogrammer. The transmitter 499 produces a digital bit stream andconverts the digital bit stream into an analog waveform (input signal532) that is modulated to the carrier frequency, e.g., 175 kHz, bymodulator 520 to produce wireless signal 533 for transmission over awireless channel. The wireless channel, in this case, is the path of thewireless signal 533 between the programming device and the IMD implantedwithin the patient.

Wireless signal 533 is received by receive antenna 502. Mixer amplifier14 receives a signal 525 from summing node 522. As previously describedwith respect to FIGS. 2, 10 and 12, mixer amplifier 14 may include anamplifier 26, a demodulator 28, and an integrator 30. Components withsimilar numbers in each of these figures may operate in a similarmanner. For example, amplifier 26 amplifies input signal 525 to producean amplified signal, i.e., amplified signal 527. Modulator 28demodulates amplified signal 527 at the chop frequency to producedemodulated signal 529, which carries the original data stream locatedback at baseband and noise modulated up to 175 kHz. Integrator 30suppresses the signal components that are out of band with the basebandcomponents, thereby producing output signal 531 which is substantiallyfree of noise 523.

As previously described with respect to FIG. 10, negative feedback path90 provides negative feedback that keeps the signal change at the inputto mixer amplifier 14 small. In particular, negative feedback path 90includes modulator 34 which modulates output signal 531 to produce adifferential feedback signal that is added to the signal path at summingnode 522. Clock signal 521C drives modulator 34 to modulate outputsignal 531 at the chopping carrier frequency via feedback capacitor 17(Cfb). Negative feedback path 90 may include two feedback path branchesthat apply the negative feedback to the positive and negative inputterminals of differential mixer amplifier 14. The feedback paths are outof phase with each other to ensure that a negative feedback path existsduring each half of the clock cycle. In this way, mixer amplifier 14provides a stable, low noise output while operating at low power.

In FIG. 14A, however, the clocks that provide clock signals 521A and521B are not located in the same physical location. In particular, clocksignal 521A is provided by a clock located in the transmitter 498 andclock signal 521B is located in instrumentation amplifier 500 inreceiver 499. Accordingly, clock signal 521B may not be synchronizedwith clock signal 521A. The phase shift between clock signals 521A, 521Bmay result in a signal null in demodulated signal 529 when the shift is90 degrees, or a beat frequency that makes decoding the received signalvery difficult if not impossible. Clock synchronizer 503 corrects forthe phase mismatch between clock signals 521A, 521B.

As shown in FIG. 14A, clock synchronizer 502 uses the received signal,i.e., input signal 533 to correct for phase mismatch between clocksignals 521A and 521B. Clock signal 521B is used by modulator 528 tochop amplified signal 527, and by modulator 34 in feedback path 90 tochop output signal 531 for feedback to summing node 522. With clocksignal 521B and clock signal 521A substantially synchronized with eachother, decoder 504 can produce a digital bitstream from output signal531. Decoder 504 may be a slicer or similar component that can convertan analog baseband signal into a digital bitstream. For example, decoder504 may include a slicer formed from a comparator that detects a levelof the output signal. The comparator may have a dynamic level adjustmentto account for variations in the background noise floor. Mild hysteresismay be added to the slicer to prevent multiple triggers in the digitalwaveform for small amplitude transitions over short periods of time.

Clock synchronizer 502 may be implemented as a phase lock loop or othercomponent known in the radio frequency (RF) communication arts thatcorrects for a phase mismatch between the clocks at the transmitter andreceiver. In one example embodiment, clock synchronizer 502 may includea chopper stabilized mixer amplifier as described in this disclosure.The chopper stabilized mixer amplifier can be used to derive the mixerclock, the clock that provides clock signal 521B to mixer amplifier 14,from the received signal thereby eliminating the need for quadraturereconstruction. In other words, the core feature of the instrumentationamplifier described in this disclosure can be used as a key buildingblock in clock synchronizer 502 for building a synchronous clock derivedfrom the received signal. This core feature has been described in detailwith respect to mixer amplifier 14 with negative feedback 90.

Using a chopper stabilized mixer amplifier in clock synchronizer 502 toderive the clock signal may have several advantages. First, the mixeramplifier is chopper stabilized, providing minimal referred to theantenna offset (RTAO). This provides a clean signal for extracting thesmall amplitude received signals, which may be on the order of 100microvolts. The use of feedback path 90 and a compensation networkallows loop dynamics to be adjusted to suppress out-of-band transientswhile maintaining lock on the received signal. In addition, signalprocessing is achieved with the chopper mixer elements, which keepcurrent drawn from a power supply to a minimum. For example, the netstandby current for instrumentation amplifier 500 with no polling may beapproximately 5 μA or less in some embodiments.

In summary, receiver 498 may have three major building blocks. The frontend at antenna 503 is attached to two chopper stabilized mixers, one ofwhich is used in a phase-lock loop 502 to derive the reference clock,and the other of which is used in mixer amplifier 14 to translate thereceived signal to baseband, and amplify it while suppressingout-of-band aggressors. In general, a chopper-stabilized mixer amplifieris provided in clock synchronizer 502 as a linear mixer to operate as aphase detector in a voltage controlled oscillator (VCO), while the otherchopper-stabilized mixer amplifier operates as a linear mixer to providedemodulation, amplification, and lowpass filtering for data extraction.The output of the in-phase mixer amplifier 14 is passed to decoder 504for digitization. The architecture of FIG. 14A provides a synchronousdemodulator that may be capable of high sensitivity to received signalsin the transmission band while rejecting out-of-band aggressors.Low-power synchronous demodulation is made possible by thechopper-stabilized mixer architecture, which may be used in mixeramplifier 14 and clock synchronizer 502.

FIG. 14B is a circuit diagram illustrating input and feedback circuitryfor the telemetry-configured instrumentation amplifier of FIG. 14A. Asshown in FIG. 14B, mixer amplifier 14 receives a modulated differentialinput signal via input capacitors 106A, 106B (Cin). Input capacitor 106Afeeds a positive end of the differential antenna signal (ANT+) to thepositive input of mixer amplifier 14. Input capacitor 106B feeds anegative end of the differential antenna signal (ANT−) to the negativeinput of mixer amplifier 14. Resistors 108A, 108B may be provided to setthe inputs of mixer amplifier 14 to set an input bias impedance.Positive and negative inputs of mixer amplifier 14 may be coupled tofeedback path branches of feedback path 90 via feedback capacitors 112A,112B (Cfb) and switches 114A, 114B, as in other embodiments. Thecapacitance of the feedback capacitor 112 (Cfb) in relation to thecapacitance of the input capacitor 106 (Cin) sets the nominal gain ofthe overall instrumentation amplifier. As in other embodiments, negativefeedback path 92 also may be provided to set a highpass cutoff for theinstrumentation amplifier.

FIG. 15A is a block diagram illustrating instrumentation amplifier 500.In accordance with this disclosure, instrumentation amplifier 500 isillustrated in FIG. 15A as including mixer amplifier 14 and feedbackpath 16. Unlike the previously described embodiments, however, front end12 is in a different physical location, consistent with FIGS. 14A and14B. In particular, as described with reference to FIG. 14A, front end12 resides within a transmitter 499 in a remote IMD or programmer. Thesignal received by receive antenna 503 of instrumentation amplifier 500has already been chopped at the remote IMD or programmer.Instrumentation amplifier 500 includes clock synchronizer 502 whichcorrects for the phase mismatch between the clock that drives front end12 in the remote device and the clock that drives mixer amplifier 14.Clock synchronizer 502 provides a linear mixer that extracts the phasereference for use in the data demodulation path provided by mixeramplifier 14.

As shown in FIG. 15A, receive antenna 503 receives the wireless signaloutput by the remote transmitter. Mixer amplifier 14 of instrumentationamplifier 500 operates as previously described and may be implemented asa modified folded cascode amplifier with switching at low impedancenodes. Thus, mixer amplifier 14 is illustrated in FIG. 15A as includingamplifier 26, demodulator 28, and integrator 30. In FIG. 15A, mixeramplifier 14 receives modulated input signal 525 from receive antenna503. Amplifier 26 amplifies modulated input signal 525 to produceamplified signal 527. Demodulator 28 demodulates amplified signal 527 toproduce demodulated signal 529 using switching at low impedance nodes ofthe folded cascode amplifier. However, demodulated signal 529 mayexperience signal nulls or a beat frequency unless the clock drivingdemodulator 28 is synchronized with the clock driving the modulator atthe transmitter. This is the reason that instrumentation amplifierincludes clock synchronizer 502.

Demodulated signal 29 may contain 1/f noise, popcorn noise, and offsetat the carrier frequency (175 kHz) and the original signal content atbaseband. Integrator 30 integrates demodulated signal 529 to produceoutput signal 531. In particular, integrator 30 integrates demodulatedsignal 529 with respect to a reference voltage provided by a receiverreference and bias generator and acts as a low pass filter to suppresssignal components with a frequency outside of the baseband.Consequently, noise sitting at the carrier frequency of demodulatedsignal 529 is substantially eliminated to produce a stable, low noiseoutput signal 531.

Again, output signal 531 is stable because of the negative feedbackprovided by negative feedback path 90. Without negative feedback path90, output signal 531 includes a series of spikes superimposed on thedesired signal that make it very difficult to slice the signal into adigital bitstream and decode the data. These spikes are a result ofoperating with very low power which limits the bandwidth of mixeramplifier 14. Providing negative feedback at the input to mixeramplifier 14 keeps the signal change small in steady state so that theonly significant voltage transitions occur at switching nodes. Negativefeedback path 90 includes symmetrical feedback path branches to providethe negative feedback to respective positive and negative differentialinputs of mixer amplifier 14. Each feedback path branch modulates outputsignal 531 with a reference voltage provided by a receiver bias andreference voltage generator. The feedback path branches are 180 degreesout of phase with each other provide feedback during each half of theclock cycle. In this way, mixer amplifier 14 and negative feedback path90 substantially eliminate glitching to provide stable, low noise outputsignal 531.

Output signal 531 may experience signal nulls or a beat frequency if thetransmitter clock and receiver clock are not in phase with each other.The transmitter clock signal drives the modulator that modulates thebaseband signal to the carrier frequency, e.g., 175 kHz. The receiverclock supplies a clock signal to mixer amplifier 14 and negativefeedback path 90. More specifically, the receiver clock supplies theclock signal that drives demodulator 28 to demodulate the received,amplified signal 527 and the signal(s) that drive modulation of theoutput signal 531 in negative feedback path 90.

Clock synchronizer 502 corrects for the phase mismatch between thetransmitter clock and the receiver clock. In particular, clocksynchronizer builds a synchronous clock derived from the receivedsignal, i.e., modulated input signal 525, to produce a correction signalthat is used by demodulator 28 in mixer amplifier 14 and the modulatorin negative feedback path 90 to compensate for the phase mismatch.

Clock synchronizer 502 in FIG. 15A avoids problems that may beassociated with using a comparator to derive the mixer clock from thereceived signal. The problems associated with using a comparator mayinclude difficulty producing a square wave because of the low powerreceived signal. That is, it may be difficult for a comparator to squareup millivolt signals at the 175 kHz clock frequency. The comparator alsotypically requires an AC coupled preamplifier or other mechanism forremoving DC offsets on the front-end, which would otherwise lead to asignificant duty cycle error and/or dead zone for signals on the orderof millivolts or less. Further, a comparator has no memory and,therefore, any signal crossing results in the signal mixing into thebaseband. This is a problem with signals on the order of hundreds ofmillivolts and, more particularly, signals on the order of hundreds ofmicrovolts with sensitivity at the 175 kHz ISM band.

In FIG. 15A, clock synchronizer 502 operates as a phase lock loop andincludes chopper stabilized mixer amplifier 560, compensation network562, voltage controlled oscillator (VCO) 564, and delay units 566 and568. Mixer amplifier 560 includes a mixer amplifier, arranged in amanner similar or identical to mixer amplifier 14. Instead of receivingnegative feedback to the inputs of the mixer amplifier, however, mixeramplifier 560 receives a quadrature phase clock feedback that is appliedto a demodulator in mixer amplifier 560. Hence, in some embodiments,chopper stabilized mixer amplifier 560 may include similar componentsand operate similar to mixer amplifier 14 described in this disclosure.For example, chopper stabilized mixer amplifier 560 may include, withrespect to FIG. 15A, an amplifier, a demodulator, and an integrator thatform a mixer amplifier and be coupled to receive a negative feedbackpath that provides the chopper stabilization for producing a stableoutput. As mentioned above, however, the negative feedback received bymixer amplifier 560 may be a quadrature phase feedback to adjust theclock frequency of the demodulator. The quadrature phase feedback is outof phase with the input signal received by mixer amplifier 560. Thus,chopper stabilized mixer amplifier 560 includes a mixer amplifier havingthe modified folded cascode amplifier architecture with switching at lowimpedance nodes. This architecture is illustrated in FIG. 6. Chopperstabilized mixer amplifier 560 is illustrated as a single block in FIG.15A.

In general, clock synchronizer 502 provides a feedback path between itsoutput and demodulator 28 of mixer amplifier 14. Chopper stabilizedmixer amplifier 560 receives modulated input signal 525 from receiveantenna 503 and produces a stable, low noise signal. Importantly,chopper stabilized mixer amplifier 560 substantially removes offset fromthe received signal and outputs a signal that substantially or closelyapproximates a square wave. As a result, chopper stabilized mixeramplifier 560 may avoid the previously discussed problems associatedwith using a comparator.

Compensation network 562 receives the output of chopper stabilized mixeramplifier 560 and applies an integrator and high-pass zero. By using anintegrator in compensation network 562, the output adjusts VCO 564 suchthat the feedback clock (output of VCO 564) is in quadrature with thereceived signal. In other words, because zero net signal is output bychopper stabilized mixer amplifier 560 in steady state, the transmitterclock and the output of VCO 564 are in quadrature. The key is that byusing an integrator in compensation network 562, the integrator holdsthe VCO value while the received signal is in the “off state” (output ofchopper stabilized mixer amplifier 560 still zero since signal is gone),and reacquires the VCO quickly when the signal goes high again. In thisway, clock synchronizer 502 can be viewed as a “phasor fly wheel” thatis locked onto the received signal, i.e., modulated input signal 25, inquadrature.

VCO 564 may operate at approximately 350 kHz (2*175 kHz ISM frequency)for the purpose of this example embodiment. The output of VCO 564 isprocessed by delay units 566 and 568 to provide quadrature signals tothe chopper stabilized mixer amplifier 560, demodulator 28, and thedemodulator in negative feedback path 90. Delay unit 568 feeds theoutput of VCO 564 back to the demodulator of chopper stabilized mixeramplifier 560. Delay unit 566 is tied to the opposite phase of VCO 564to create an in-phase clock for demodulator 28 and negative feedbackpath 90. That is, because the output of VCO 564 is locked onto the inputsignal in quadrature, delay unit 566 introduces delay of half a clockcycle to create an in-phase clock for mixer amplifier 14 (demodulator28) and negative feedback path 90. Hence, delay unit 566 is configuredto feed the output of VCO 564 with a first phase (d) to demodulator 28of mixer amplifier 14 and modulator 34 of negative feedback path 90,while delay unit 568 is configured to feed the output of VCO 564 with asecond phase of Φ′ to the demodulator in mixer amplifier 560. Theoutputs of delay units 566 and 568 are 90 degrees out of phase with oneanother. With demodulator 28 using a clock signal that is in phase withthe transmitter clock, signal processing can be applied to the output ofmixer amplifier 14 to recover and decode the transmitted bits. Delayunits 566 and 568 may be D-type flip flops or other components that canbe used to introduce delay into the signal.

In general, clock synchronizer 502 may be a phase-locked-loop thatextracts the phase reference for the data demodulation path of mixeramplifier 14 and negative feedback path 90. The feedback from VCO 564adjusts the modulation clock of chopper stabilized mixer amplifier 560such that it is 90 degrees out of phase with the clock frequency of theinput signal 525. In this case, chopper stabilized mixer amplifier 560may act as a linear phase detector having an output that scales asVin*cos(Φ), where Vin is the input voltage from receive antenna 503, andΦ is the phase difference between the chopper stabilized mixer amplifier560 and the input signal. The resulting transfer function has a null at90 degrees. For purposes of feedback compensation, small variationsabout that point can be approximated as a linear relationship.

The compensation of VCO 560 by compensation network 562 may becomplicated by the fact that loop gain scales with the input voltage. Byusing a simple integrator with a zero in compensation network 562, astable phase lock can be obtained for small signals at the antenna 503.For large voltages, however, the compensation zero creates a largesignal at the clock frequency that can saturate the channel and throwoff the VCO. The origin of this signal is the “hidden state” of themixer output at lock, which has no DC component, but a significantsignal at the mixer frequency. To eliminate this problem, a second polecan be added to the compensation network 562 beyond loop cross-over. Thepurpose of this pole is to suppress the signal at the mixer frequencyand minimize VCO jitter. As long as the loop gain is not too high, theadditional pole should not be a problem. The extra pole is pulled intothe compensation zero, which acts to pull the double integrator (onefrom the mixer amplifier 560 and one from the phase integration of VCO564) off the imaginary axis and into the left half plane.

As VCO 564 is carefully compensated, a robust lock can be achievedacross the dynamic range of the telemetry link. In this way, the loopmay be optimally compensated in that it responds more slowly andtherefore more heavily filters disturbances as the telemetry linkdistance increases and signals fall off. In practice, a mode switchdriven by received signal strength (RSSI) may be provided to maintainsomewhat uniform dynamics over a typical telemetry link range. The modeswitch may operate to adjust the loop gain of clock synchronizer 502based on the level of the input signal. Hence, the loop gain may bedecreased for higher input signal levels and increased for lower inputsignal levels.

FIG. 15B is a block diagram illustrating a clock synchronizer 502 inFIG. 15A in greater detail. FIG. 15B illustrates clock synchronizer 502substantially as shown in FIG. 15A, but further illustrates examplecomponents of mixer amplifier 560. In particular, mixer amplifier 560may include amplifier 26B, modulator 28B, and integrator 30B, all ofwhich may function in a manner similar to amplifier 26, modulator 28 andintegrator 30 of mixer amplifier 14. As shown in FIG. 15B, however,delay unit 568 feeds the output of VCO 564 in quadrature phase withinput signal 525 to adjust modulator 28B of mixer amplifier 560. Hence,the feedback signal for modulator 28B is 90 degrees out of phase withthe input signal 525, and is used to adjust the clock frequency ofmodulator 28B and thereby maintain chopper stabilization of mixeramplifier 560.

FIG. 16 is a block diagram illustrating various components of animplantable medical device (IMD) 700 including an instrumentationamplifier as described in this disclosure. IMD 700 includes therapydelivery module 702, processor 704, memory 708, telemetry module 706,sensor 710, power source 712, and therapy elements 714. In general, IMD700 includes a chopper stabilized instrumentation amplifier as part ofsensor 710, telemetry module 76, or both.

Sensor 710 may be a pressure sensor, accelerometer, activity sensor,impedance sensor, electrical signal sensor or other sensor configured tomonitor heart sounds, brain signals, and/or other physiological signals.Although illustrated in FIG. 16 as contained within IMD 700, a portionof sensor 710 may be located outside of IMD 700. For example, a sensortransducer or one or more electrodes may be located on a distal tip of alead implanted at a target site within the patient and electricallycoupled to IMD 700 via conductors. Alternatively, a sensor transducer orone or more electrodes may be provided on or within a housing of IMD700. For example, an accelerometer may be provided within an IMD housingor within a lead that extends from the IMD. To sense electrical signals,sensor 710 may include two or more electrodes arranged on a lead, anelectrode on a lead and an electrode on an IMD housing, two or moreelectrodes arranged on an IMD housing, or other electrode arrangements.Sensor circuitry associated with sensor 710 may be provided withinsensor 710 in the housing of IMD 700.

In general, sensor 710 provides a measurement of a physiological signalor parameter by translating signal or parameter to an output voltage orcurrent. A chopper stabilized instrumentation amplifier amplifies andfilters the sensor output as described in this disclosure to produce astable, low noise signal with very low power requirements. In this way,the chopper stabilized instrumentation amplifier may enable IMD 700 tooperate for several months or years relying on power from a finite powersource 712, such as a rechargeable or nonrechargeable battery. In eithercase, power conversation is desirable.

The output of sensor 710 and, more particularly, the output of thechopper stabilized instrumentation amplifier associated with sensor 710may be received by processor 704. Processor 704 may apply additionalprocessing, e.g., convert the output to digital values for processing,prior to storing the values in memory 708, and/or transmitting thevalues to an external programmer via telemetry module 706. Telemetrymodule 706 also may include at least a portion of a chopper-stabilizedinstrumentation amplifier. Processor 704 may also control delivery oftherapy to the patient based on the output of sensor 710.

IMD 700 may deliver therapy to a patient via therapy elements 714. Inother embodiments, IMD 700 may be dedicated to sensing and may notinclude therapy delivery module 702. Therapy delivery elements 714 maybe electrodes carried on one or more leads, electrodes on the housing ofIMD 700, one or more fluid delivery devices, or any combination thereof.Accordingly, therapy delivery module 702 may include an implantablestimulation generator or other stimulation circuitry that deliverselectrical signals, e.g., pulses or substantially continuous signals,such as sinusoidal signals, to the patient via at least some of theelectrodes that form therapy elements 714 under the control of processor704.

The stimulation energy generated by therapy delivery module 40 may beformulated as stimulation energy for treatment of any of a variety ofcardiac or neurological disorders, or disorders influenced by patientneurological response. Example stimulation therapies include cardiacpacing, cardiac defibrillation, deep brain stimulation (DBS), spinalcord stimulation (SCS), peripheral nerve field stimulation (PNFS),pelvic floor stimulation, gastrointestinal stimulation, and the like.

Therapy delivery module 702, processor 704, telemetry module 706, memory708, and sensor 710 receive operating power from power source 712. Powersource 712 may take the form of a small, rechargeable ornon-rechargeable battery, or an inductive power interface thattranscutaneously receives inductively coupled energy. In the case of arechargeable battery, power source 712 similarly may include aninductive power interface for transcutaneous transfer of recharge power.

In embodiments in which one or more fluid delivery devices are part oftherapy elements 714, therapy delivery module 702 may include a one ormore fluid reservoirs and one or more pump units that pump fluid fromthe fluid reservoirs to the target site through the fluid deliverydevices. The fluid reservoirs may contain a drug or mixture of drugs.The fluid reservoirs may provide access for filling, e.g., bypercutaneous injection of fluid via a self-sealing injection port. Thefluid delivery devices may comprise, for example, catheters thatdeliver, i.e., infuse or disperse, drugs from the fluid reservoirs tothe same or different target sites.

Processor 704 may include a microprocessor, microcontroller, digitalsignal processor (DSP), application specific integrated circuit (ASIC),field programmable gate array (FPGA), discrete logic circuitry, or acombination of such components. Processor 704 is programmed to controldelivery of therapy according to a selected parameter set stored inmemory 708. Specifically, processor 704 controls therapy delivery module702 to deliver electrical stimulation, drug therapy, or a combination ofboth. For example, processor 704 may control which drugs are deliveredand the dosage of the drugs delivered.

Processor 704 may also control therapy delivery module 702 to deliverelectrical stimulation with pulse amplitudes, pulse widths, andfrequencies (i.e., pulse rates) specified by the programs of theselected parameter set. Processor 704 may also control therapy deliverymodule to deliver each pulse according to a different program of theparameter set. In some embodiments, processor 704 may control therapydelivery module 702 to deliver a substantially continuous stimulationwaveform rather than pulsed stimulation.

Memory 708 may store parameter sets that are available to be selected bythe patient for delivery of electrical stimulation and/or drug therapy.Memory 42 may also store schedules. Memory 708 may include anycombination of volatile, non-volatile, removable, magnetic, optical, orsolid state media, such as read-only memory (ROM), random access memory(RAM), electronically-erasable programmable ROM (EEPROM), flash memory,or the like.

Processor 704 also controls telemetry module 706 to exchange informationwith an external programmer, such as a clinician programmer and/orpatient programmer by wireless telemetry. Processor 704 may controltelemetry module 706 to communicate with the external programmer on acontinuous basis, at periodic intervals, or upon request from theprogrammer. In addition, in some embodiments, telemetry module 706 maysupport wireless communication with one or more wireless sensors thatsense physiological signals and transmit the signals to IMD 700.

Telemetry module 706 may operate as a transceiver that receivestelemetry signals from an external programmer and transmits telemetrysignals to an external programmer. In some embodiments, telemetry module706 may include a chopper stabilized instrumentation amplifier. Morespecifically, with respect to FIGS. 14 and 15, the receiver portion oftelemetry module 706 may include the back end of a chopper stabilizedinstrumentation amplifier, referred to as a chopper stabilized mixeramplifier and feedback path, that produces a baseband signal from areceived telemetry signal. The receiver portion is described in thisdisclosure as including only the back end (chopper stabilized mixeramplifier) because the corresponding front end is located in thetransmitter portion of the external programmer in communication with IMD700.

The receiver portion may also include a clock synchronizer that includesanother chopper stabilized mixer amplifier, e.g., as described withreference to FIG. 15A. This chopper stabilized mixer amplifier producesan output that can be used by a phase lock loop to generate a correctionsignal that is used to synchronize the receiver portion of telemetrymodule 706 with the transmitter of the external programmer.

Telemetry module 706 also may include a transmitter to transmit signalsfrom IMD 700 to an external programmer or to another IMD or externalmedical device. The transmitter may include a front end of achopper-stabilized instrumentation amplifier in the sense that it mayinclude a first chopper stage that modulates an input signal with an RFfrequency for transmission to an external programmer or anotherimplanted or external medical device.

Importantly, the instrumentation amplifiers in sensor 710 and telemetrymodule 706 are micropower circuits that provide stable, low noisesignals. Thus, IMD 700 may operate over a longer duration of time thanwould be possible using instrumentation amplifiers that require morepower for operation.

FIG. 17 is a block diagram illustrating an example patient or clinicianprogrammer 720 that allows a patient or clinician to communicate withIMD 700. A patient or clinician may interact with programmer 720 toprogram therapy, e.g., electrical stimulation, drug therapy, or acombination of both. In the illustrated example, programmer 720 includesprocessor 722, user interface 724, input/output 726, telemetry module728, memory 730, and power source 732. Programmer 720 may include achopper stabilized instrumentation amplifier as part of telemetry module728.

A patient or clinician, referred to as a user herein, may interact withprocessor 722 via user interface 724 in order to control delivery ofelectrical stimulation, drug therapy, or a combination of both. Userinterface 724 may include a display and a keypad, and may also include atouch screen or peripheral pointing devices as described above.Processor 722 may also provide a graphical user interface (GUI) tofacilitate interaction with the user, as will be described in greaterdetail below. Processor 722 may include a microprocessor, a controller,a DSP, an ASIC, an FPGA, discrete logic circuitry, or the like.

Programmer 720 also includes memory 730. In some embodiments, memory 730may store parameter sets that are available to be selected by the userfor delivery of therapy. Memory 730 may also store schedules. Hence,parameter sets and schedules may be stored in IMD 700, programmer 720,or both. Programmer 720 also includes a telemetry module 728 that allowsprocessor 722 to communicate with IMD 700, and, optionally, input/outputcircuitry module 726 that allows processor 722 to communicate withanother programmer.

Processor 722 may receive parameter set selections made by the user viauser interface 724, and may either transmit the selection or theselected parameter set to IMD 700 via telemetry circuitry 728 to delivertherapy according to the selected parameter set. Where programmer 720stores parameter sets in memory 730, processor 722 may receive parametersets from another programmer via input/output module 726 duringprogramming by a clinician. For example, a patient programmer mayreceive parameter sets from a clinician programmer.

Telemetry module 728 may include a transceiver for wirelesscommunication, appropriate ports for wired communication orcommunication via removable electrical media, or appropriate drives forcommunication via removable magnetic or optical media. If wirelesscommunication is used, telemetry module 728 may support both wirelesscommunication with IMD 700 and wireless communication with anotherprogrammer.

Similar to telemetry module 706 of IMD 700, telemetry module 728operates as a transceiver for transmitting and receiving signals to andfrom IMD 700 and possibly another programmer. The receiver portion oftelemetry module 728 may include a chopper stabilized mixer amplifier inthe main signal path for producing a baseband signal that can beprocessed to recover the transmitted signal. The corresponding front endto this chopper stabilized mixer amplifier is located in the transmitterportion of IMD 700.

The receiver portion may also include a chopper stabilized mixeramplifier in a clock synchronizer or phase lock loop for the main signalpath. This chopper stabilized mixer amplifier down mixes the receivedsignal to baseband to produce a signal that is processed by the phaselock loop to derive a synchronous clock. The transmitter portion oftelemetry module 728 may include a first chopper stage that chops aninput signal at an RF frequency for transmission to IMD 700 or otherprogrammers or devices.

Power source 732 provides power to programmer 720. That is, power source732 provides power to processor 722, user interface 724, input/outputmodule 726, telemetry module 728, and memory 730. Because chopperstabilized mixer amplifiers in telemetry module 728 operate at very lowpower, they may increase the life of power source 732.

Power source 732 may take the form of a small, rechargeable ornon-rechargeable battery, or an inductive power interface thattranscutaneously receives inductively coupled energy. In the case of arechargeable battery, power source 732 similarly may include aninductive power interface for transcutaneous transfer of recharge power.

The invention, including instrumentation amplifiers and associatedcircuitry, devices, systems and methods, may be useful in a variety ofapplications, For example, the invention may be applied to supportsensing relating to therapies for a variety of symptoms or conditionssuch as cardiac arrhythmia, cardiac fibrillation, chronic pain, tremor,Parkinson's disease, epilepsy, urinary or fecal incontinence, sexualdysfunction, obesity, or gastroparesis, and may provide informationuseful in controlling electrical stimulation or drug delivery to avariety of tissue sites, such as the heart, the brain, the spinal cord,pelvic nerves, peripheral nerves, or the gastrointestinal tract of apatient.

Hence, an instrumentation amplifier as described in this disclosure maybe integrated with, housed in, coupled to, or otherwise associated withan external or implantable medical device, such as acardioverter/defibrillator, spinal cord stimulator, pelvic nervestimulator, deep brain stimulator, gastrointestinal stimulator,peripheral nerve stimulator, or muscle stimulator, and also may be usedin conjunction with implantable or external drug delivery devices. Forexample, an instrumentation amplifier and/or associated sensing devicesmay reside within an implantable medical device housing or a lead orcatheter coupled to such a device.

The instrumentation amplifier may be used in conjunction with differenttherapeutic applications, such as cardiac stimulation, deep brainstimulation (DBS), spinal cord stimulation (SCS), pelvic stimulation forpelvic pain, incontinence, or sexual dysfunction, gastric stimulationfor gastroparesis, obesity or other disorders, or peripheral nervestimulation for pain management. Stimulation also may be used for musclestimulation, e.g., functional electrical stimulation (FES) to promotemuscle movement or prevent atrophy.

Various embodiments of the invention have been described. These andother embodiments are within the scope of the following claims.

1. An electrical impedance sensing device comprising: a current sourceconfigured to generate a modulated current at a modulation frequency forapplication to a load to produce an input signal; an amplifierconfigured to amplify the input signal to produce an amplified signal; ademodulator configured to demodulate the amplified signal at themodulation frequency to produce an output signal indicating an impedanceof the load.
 2. The device of claim 1, further comprising a modulatorconfigured to modulate the output signal at the modulation frequency toproduce a feedback signal.
 3. The device of claim 2, further comprisinga feedback path that applies the feedback signal to the input signal. 4.The sensing device of claim 3, further comprising an input capacitorthat couples the input signal to an input of the amplifier, wherein thefeedback path is coupled to apply the feedback signal to the inputsignal at a node between the input capacitor and the input of theamplifier via a feedback capacitor.
 5. The sensing device of claim 4,wherein the input capacitor and the feedback capacitor are arranged suchthat a ratio of the feedback capacitor to the input capacitor sets again of the amplifier.
 6. The sensing device of claim 3, wherein theamplifier includes a differential amplifier, the input signal is adifferential input signal, the output signal is a differential outputsignal, and the current source includes a differential current source.7. The sensing device of claim 6, wherein the modulator includes a firstmodulator that modulates a first component of the differential outputsignal at the modulation frequency and a second modulator that modulatesa second component of the differential output signal at the modulationfrequency, and wherein the feedback path includes a first feedback pathbranch that couples the first component of the differential outputsignal to the first input of the differential amplifier and a secondfeedback path branch that couples the second component of thedifferential output signal to the second input of the differentialamplifier.
 8. The sensing device of claim 1, wherein the current sourcecomprises a voltage source and a switch that modulates a currentproduced by the voltage source to produce the modulated current.
 9. Thesensing device of claim 1, wherein the load includes a biological load,the device further comprising first implantable electrodes coupled toapply the modulated current across a biological load, and secondimplantable electrodes coupled to sense the input signal produced acrossthe biological load.
 10. A biological impedance sensing devicecomprising: means for applying a current modulated at a modulationfrequency across a load to produce an input signal; means for amplifyingthe input signal to produce an amplified signal; means for demodulatingthe amplified signal at the modulation frequency to produce an outputsignal indicating an impedance of the load.
 11. The device of claim 10,further comprising means for modulating the output signal at themodulation frequency to produce a feedback signal.
 12. The device ofclaim 11, further comprising means for applying the feedback signal tothe input signal.
 13. The sensing device of claim 12, further comprisingan input capacitor that couples the input signal to an input of themeans for amplifying, and means for applying the feedback signal to theinput signal at a node between the input capacitor and the input of theamplifying means via a feedback capacitor.
 14. The sensing device ofclaim 13, wherein the input capacitor and the feedback capacitor arearranged such that a ratio of the feedback capacitor to the inputcapacitor sets a gain of the amplifier.
 15. The sensing device of claim12, wherein the amplifying means includes a differential amplifier, theinput signal is a differential input signal, and the output signal is adifferential output signal.
 16. The sensing device of claim 15, whereinthe modulating means includes a first modulator that modulates a firstcomponent of the differential output signal at the modulation frequencyand a second modulator that modulates a second component of thedifferential output signal at the modulation frequency, and wherein themeans for applying the feedback signal includes a first feedback pathbranch that couples the first component of the differential outputsignal to the first input of the differential amplifier and a secondfeedback path branch that couples the second component of thedifferential output signal to the second input of the differentialamplifier.
 17. The sensing device of claim 10, wherein the means forapplying the modulated current comprises a voltage source and a switchthat modulates a current produced by the voltage source to produce themodulated current.
 18. The sensing device of 10, wherein the loadincludes a biological load, the device further comprising firstimplantable electrodes coupled to apply the modulated current across abiological load, and second implantable electrodes coupled to sense theinput signal produced across the biological load.
 19. An implantablemedical device comprising: a therapy delivery module configured todeliver a therapy to a patient; an impedance sensor comprising: acurrent source configured to generate a modulated current at amodulation frequency for application across a biological load to producean input signal, an amplifier configured to amplify the input signal toproduce an amplified signal, and a demodulator configured to demodulatethe amplified signal at the modulation frequency to produce an outputsignal indicating an impedance of the biological load; and a processorconfigured to control the therapy delivery module and process arepresentation of the output signal produced by the sensor.
 20. Thedevice of claim 19, further comprising a modulator configured tomodulate the output signal at the modulation frequency to produce afeedback signal.
 21. The device of claim 20, further comprising afeedback path that applies the feedback signal to the input signal. 22.The device of claim 19, wherein the therapy delivery module comprises anelectrical stimulation delivery module.
 23. The device of claim 19,wherein the therapy delivery module comprises a fluid delivery module.24. A method for sensing impedance of a load, the method comprising:applying a current modulated at a modulation frequency across a load toproduce an input signal; amplifying the input signal with an amplifierto produce an amplified signal; and demodulating an amplitude of theamplified signal at the modulation frequency to produce an output signalindicating an impedance of the load.
 25. The method of claim 24, furthercomprising modulating an amplitude of the output signal at themodulation frequency to produce a feedback signal.
 26. The method ofclaim 25, further comprising applying the feedback signal to the inputsignal via a feedback path.
 27. The method of claim 26, furthercomprising coupling the input signal to an input of the amplifier via aninput capacitor, wherein applying the a feedback signal to the inputsignal comprises applying the feedback signal to the input signal at anode between the input capacitor and the input of the amplifier via afeedback capacitor.
 28. The method of claim 27, wherein the inputcapacitor and the feedback capacitor are arranged such that a ratio ofthe feedback capacitor to the input capacitor sets a gain of theamplifier.
 29. The method of claim 26, wherein the amplifier includes adifferential amplifier, the input signal is a differential input signal,and the output signal is a differential output signal.
 30. The method ofclaim 29, further comprising modulating a first component of thedifferential output signal at the modulation frequency and modulating asecond component of the differential output signal at the modulationfrequency, applying the first component of the differential outputsignal to the first input of the differential amplifier, and applyingthe second component of the differential output signal to the secondinput of the differential amplifier.
 31. The method of claim 24, furthercomprising switching a current produced by a voltage source at themodulation frequency to produce the modulated current.
 32. The method ofclaim 24, wherein the load includes a biological load, the methodfurther comprising applying the modulated current across the biologicalload via first implantable electrodes, and sensing the input signalproduced across the biological load via second implantable electrodes.